• 专利附录
  • 专利说明
  • 权利要求
  • 类似技术
  • 同族专利
  • 引用文献
  • 法律状态
  • 优先权
    • 专利摘要:
    The inventive drive circuit for controlling a power transistor, in particular a gate-injection transistor, comprises: • an upper terminal (p) and a lower terminal (0) of a supply terminal for connection to a unipolar Ansteuerversorgungsspannungsquelle, • a first control terminal (1) and a second control terminal (2) for connection to an electronic switch to be controlled; • a coupling capacitor (C1) and a switch-on current resistor (R3). At least one of the following elements is present: a limiting element, which limits a voltage across the coupling capacitor C1 to values of only one polarity; A discharge limiting element which limits a discharge of the coupling capacitor (C1); A control voltage limiting element which limits a voltage between the first control terminal (1) and the second control terminal (2) to a maximum value (U21lim); A connection switch between the control terminals (1, 2) to charge the coupling capacitor (C1) before first turning on the power transistor without turning on the power transistor.
    公开号:CH711457A2
    申请号:CH00085/16
    申请日:2016-01-21
    公开日:2017-02-28
    发明作者:Bortis Dominik；Marco Knecht Oliver；Walter Kolar Johann
    申请人:Eth Zürich Eth Transfer；
    IPC主号:
    专利说明:

    State of the art
    For controlling electronic switching elements, e.g. Power transistors are used according to the prior art driving circuits which form to ensure low switching losses at the beginning of a switch-on or switch-off a relatively high positive or negative drive current, which then goes back in the sense of low drive losses to a low Ansteuereinschaltruhestrom, wherein a positive (turn-on interval) or negative (turn-off interval) drive voltage typically occurs between the upper and lower control terminals (eg, referred to as gate and source or base and emitter) to safely hold the device in the on or off state. For purely voltage controlled electronic switches, e.g. Power MOSFETs, the drive quiescent current in this case in the on and off interval to zero; For other electronic switches may be required due to the inner component structure in the on and / or off interval a small drive quiescent current.
    The practical realization of the drive circuit must be done in practice for space and cost reasons and in view of the reliability with minimal circuit complexity and minimal complexity. In particular, to supply the drive circuit advantageously only one voltage source, i. a unipolar feeding can be used. In order nevertheless for the switching on of the electronic switch to be able to form a positive and for switching off a negative drive voltage and beyond to achieve high dynamics of the switching on and off, according to the prior art, the output of an input side between an upper terminal p and a lower terminal 0 of a supply terminal of a drive supply voltage source U1 (the upper terminal has positive potential with respect to the lower terminal) transistor switching stage not directly but via a coupling capacitor C1 connected to the upper control terminal 1 of an electronic switch to be controlled S and the lower control terminal 2 directly to the lower terminal 0 is set by the drive supply voltage source U1, which also represents the reference potential. The transistor switching stage has e.g. an open bridge branch structure (see FIG. 1), thus consists of an upper drive transistor or switch S1 branching off from the upper terminal p of U1 in the current flow direction via an upper series resistor R1 to the output A of the transistor switching stage and thus to the upper control terminal 1 of the electronic switch facing away from the first terminal of the coupling capacitor C1 is connected. Furthermore, the output A of the transistor switching stage and thus the first terminal of the coupling capacitor C1 via a lower series resistor R4 and a lower drive transistor or switch S2 in the current flow direction to the lower terminal 0 of the Ansteuerversorgungsspannung or reference potential is connected. Finally, a switch-on current resistance R3 is applied to the lower terminal of the upper drive transistor S1 facing away from the upper terminal p of the drive supply voltage and connected to the second control terminal 1 of the electronic switch S at the second end. Alternatively, the switch-on current resistance R3 can also be connected directly in parallel with C1.
    For the further embodiments, it is assumed that for switching on the electronic switch S a sufficiently high, i. sufficiently far above the Einschaltschwellspannung (threshold voltage) Uth of S lying (physically from the upper control terminal 1 to the lower control terminal 2 facing) (outer) drive voltage uSt and for switching off a negative, or at least below the threshold voltage Uth lying drive voltage uSt apply is. Furthermore, as the drive-side internal equivalent circuit of the electronic switch S, an inner drive resistor R2 is assumed to be at the upper end M of an internal equivalent capacitance C2, the lower end of C2 being at the lower control connection 2 and at reference potential, respectively Diode D2 in the current flow direction against the lower control terminal 2 is acting acting to show (depending on the internal component structure of the switch S may be provided for the equivalent circuit, no diode D2). The conducting or blocking state of switch S is then determined by the voltage appearing at C2, i. determined by an internal control voltage uC2. If stationary uC2> Uth, the switch S is in the switched-on state, for stationary uC2 <Uth in the switched-off state. Finally, the retroactive effect of a change in the voltage between the output or power terminals LI and L2 of the electronic switch S (hereinafter referred to as switch voltage uS) is taken into account by a Millerkapazität CM lying against the upper end M of C2, at the second end of the switch voltage uS, which is thought to be measured against reference potential, attacks. The switch-on current resistance R3 is directly parallel to C1, but can also be applied to the upper control connection 1 branching off from the circuit point between S1 and R1.
    For switching on the electronic switch S S1 is turned on and charged so C2 via S1-R1-C1-R2 starting from zero voltage until the arrival of Uth switching on the switch S begins. The detailed further process, in particular the retroactive effect of uS via CM on the course of uC2 (millereffect) is of secondary importance here, it is only important that uC2 finally reaches the forward voltage UD2 of D2 and is clamped to this value by means of D2; C1 is then further charged with the time constant (R1 + R2) xC2 to the positive final value U C1e = U1-UD2. Accordingly, with a correspondingly low-impedance choice of R1 and R2, u C2 = U th is reached rapidly, i. S is switched on with only a small delay. After the end of charging of C1 stationarily flows only a relatively small Einschaltruhestrom on the relatively high-impedance resistor R3 (C2 then represents an interruption), which ensures a low on-resistance of S.
    For the switching off of S S1 is disabled and subsequently switched through S2 and thus C1 connected via R2, R4 and S2 parallel to C2 and thus C2 reloaded to C1 and C2 is the same voltage. C1 is selected in the capacitance value so that uC1 in any case maintains the positive polarity present in the switch-on interval; Ultimately, this results in a physically negative internal control voltage uC2 (as above, the detailed switch-off process of S, which starts as soon as uC2 falls below the value Uth, is not considered further). It is only to be noted that when the R4 is set to a low impedance, switching off again occurs rapidly, i. takes place with little delay and that despite the supply of the drive circuit with only one supply voltage U1 in the off interval, a negative internal control voltage uC2 can be formed. C1 as well as C2, however, are slowly discharged via the switch-on current resistor R3 (for C2 the discharge takes place via R4 - if necessary R1 - R3 - R2), so that the negative internal control voltage decreases noticeably.
    For the renewed switching of S S2 is disabled and S1 is turned on, whereby C2 recharged via the path R1, C1, R2 again to a positive voltage value uC2> Uth and so again a low-ohmic state of S is achieved. The height of the initially occurring switch-on control current is dependent on the depth of the preceding discharge of C1 via R3 in the switch-off interval. Advantageously, therefore, R3 is not chosen to high impedance to achieve a sufficient reduction in voltage even with a short switch-off. However, there is a problem with the operational safety, if uC2 (or uC1) is finally reduced to zero with a longer switch-off duration. If a steep change in the switch voltage uS then occurs within the switch-off interval, as is the case with the arrangement of electronic switches in the bridge branch configuration according to the switching of the switch located in the bridge branch, then the transfer of the Miller current flowing through the Miller capacitance CM could result in a transfer of C1 or C2 With this, uC2 ev. values uC2> Uth would be reached, which, despite the applied switch-off command, would result in a parasitic switching of S and thus a bridge short-circuit, which would typically lead to the destruction of S.
    This disadvantageous effect also occurs for an alternative embodiment of the Ansteuerschaltstufe, where the Ansteuerschaltstufe by arrangement of the drive transistors S1 and S2 in the form of a lying between p and 0 closed Ansteuerbrückenzweiges (direct series connection of S1 and S2, see Fig. 2) is formed and the output A of the switching stage is tapped between S1 and S2 and connected via a turn-on resistor R1 to C1 and branching off from this point a turn-off resistor R4 is guided via a diode D4 in the current flow direction back to A. Furthermore, the switch-on current resistance R3 is applied from A to the upper control terminal 1 or connected directly in parallel to C1. This embodiment has the same basic function as described above for the use of an open control bridge branch, and substantially the same operational safety issues.
    One way to achieve a higher reliability is to perform the supply of the drive circuit bipolar, i. to provide a second supply voltage source U2, which is connected to the positive terminal to the negative terminal 0 of U1 or reference potential. The lower drive transistor S2 previously connected to reference potential is then switched to the negative terminal n of U2 for open or closed bridge branch structure instead of reference potential. Even with discharged capacitor C1, i. For μC1 = 0 (e.g., at start of operation) then during the turn-off state of S, i. In the switch-on interval of S2 stationary, the negative voltage of -U2 at control terminal 1, whereby a higher voltage reserve exists, before a negative charging of C1 by a Miller current causes a turn on of S. However, the implementation cost of the drive circuit is significantly increased by the additional supply voltage U2.
    The object of the invention is therefore to extend the state of the art unipolar drive circuits with a view to maintaining low complexity such that regardless of the duration of the off state or the duty cycle (relative duration of the off interval based on the total duration of a and off cycle)<tb> • <SEP> is present before restarting a defined voltage uC1 and / or<tb> • <SEP> the coupling capacitor C1 can not be completely discharged at least and / or<tb> • <SEP> a Miller current occurring in the turn-off interval can not cause C1 to be reloaded.
    In the sense of broad applicability, all modifications except for unipolar and bipolar supply of the drive circuit should be used.
    The object is achieved by at least one of the circuits according to the claims.
    The drive circuit for driving a power transistor, in particular a gate-injection transistor, thus has:<tb> • <SEP> an upper terminal and a lower terminal of a supply terminal for connection to a unipolar drive supply voltage source,<Tb> <September><tb> • <SEP> a first control terminal and a second control terminal for connection to an electronic switch to be controlled;<tb> • <SEP> a coupling capacitor and a switching stage with electronic switches.
    In a first circuit variant<tb> • <SEP> is by means of the switching stage, a first current path from the first control terminal through the coupling capacitor optionally formed to the upper terminal or the lower terminal,<tb> • <SEP> and the drive circuit has a switch-on current resistor which is arranged in a second current path from the first control connection to the same terminal of the supply connection to which the first current path leads in each case and parallel to the coupling capacitor.
    In a second circuit variant<tb> • <SEP> is by means of the switching stage, a first current path from the first control terminal (1) selectively to the upper terminal (p) or lower terminal (0) formed, and is the coupling capacitor (C1) between the second control terminal (2) and the lower terminal (0),<tb> • <SEP> and the drive circuit has a switch-on current resistor (R3), which is arranged in a second current path from the second control connection (2) to the lower terminal (0) of the supply connection, and parallel to the coupling capacitor (C1),
    At least one of the following elements is present, which applies to both circuit variants:<tb> • <SEP> a limiting element which limits a voltage across the coupling capacitor to values of only one polarity;<tb> • <SEP> a discharge limiting element which limits a discharge of the coupling capacitor by the Einschaltruhestromwiderstand to a minimum voltage;<tb> • <SEP> a control voltage limiting element that limits a voltage between the first control terminal and the second control terminal measured from the second to the first control terminal to a maximum value;<tb> • <SEP> a connection switch between the first control terminal and the second control terminal, with an auxiliary controller which is adapted to turn on the connection switch before a first turn on the power transistor to charge the coupling capacitor without the power transistor is thereby turned on.
    The limiting element thus limits the voltage across the coupling capacitor to only positive or negative values, depending on how the polarity of the voltage is defined and optionally depending on the type of power transistor (p or n-type). If, for example, a positive voltage is applied to the coupling capacitor for switching on the power transistor, then the limiting element limits the voltage to positive values.
    The first and the second current path thus lead both from the first control terminal to either the upper terminal or the lower terminal. You can perform sections of the same elements, but always lead the coupling capacitor and the Einschaltruhestromwiderstand parallel current paths. Thus, the coupling capacitor allows a short and relatively high current and allows the Einschaltruhestromwiderstand a continuous and relatively small current between the first control terminal and the respective terminal.
    The circuit is particularly suitable for the control of normal-off GaN gate injection transistors.
    In one embodiment, the limiting element is a diode, in particular a Schottky diode.
    In one embodiment, the discharge limiting element is a diode which blocks a discharge of the coupling capacitor by the Einschaltruhestromwiderstand.
    This diode may, in the open design of a Ansteuerbrückenzweiges be a switched in the first current path and not in the second current path diode, which does not allow a current flow from the upper terminal to the coupling capacitor but vice versa.
    This diode, if coupling capacitor and Einschaltruhestromwiderstand are connected in parallel, be a directly connected in series to Einschaltruhestromwiderstand zener diode.
    In one embodiment, the control voltage limiting element is a series circuit of a Zener diode and another diode.
    In one embodiment, the additional control is adapted to turn off the connection switch when a sufficiently high voltage is established at the coupling capacitor
    In the following, the subject invention based on preferred embodiments, which are illustrated in the accompanying drawings, explained in more detail. Each show schematically:<Tb> FIG. 1: <SEP> First basic form of a drive circuit with unipolar supply U1 according to the prior art with execution of the Ansteuschaltstufe in the form of an open bridge branch; shown further: time course of the voltage uC1 of the coupling capacitor C1 and the inner control voltage uC2 of the switch S and the control signals of the transistors S1 and S2 of the transistor switching stage.<Tb> FIG. 2: <SEP> Second basic form of a drive circuit with unipolar supply U1 according to the prior art with execution of the Ansteuerschaltstufe in the form of a closed bridge branch.<Tb> FIG. 3: <SEP> Extensions of the Circuit of FIG. 1; shown further: time course of the voltage uC1 of the coupling capacitor C1 and the inner control voltage uC2 of the switch S and the control signals of the transistors S1 and S2 of the transistor switching stage.<Tb> FIG. 4: <SEP> Extensions of the circuit according to FIG. 2.<Tb> FIG. 5 <SEP> Modification of the drive circuit according to FIG. 4 and associated characteristic time courses of control signals and voltages. As a result of the modification, a negative internal drive voltage uC2 is already ensured before the first switch-on of the electronic switch S at the time t2.
    The circuit extensions 1-4 described below may be present individually or in combination:
    1. In order to prevent the reloading of C1 by a Miller current within the turn-off interval, a circuit path is inserted parallel to C1, which prevents a reversal of the polarity of the voltage uC1.In the simplest case, this parallel path can be realized by a low-pass clamping diode (for example a Schottky diode), which is laid parallel to C1 from the upper control connection 1 in the current flow direction, as shown in FIGS. 3 and 4. Since C1 also has a positive voltage during regular operation in the switch-off interval, which represents a blocking voltage for DK, DK is not conducting current during regular operation and therefore has no influence on the operation of the control circuit. An advantage of DK is given in particular at the first switching on, arranged in a bridge branch structure electronic switches. C1 is then still uncharged and switching on the opposite in the bridge branch switch would possibly cause a negative charge of C1 without DK, which could lead to parasitic switching on the switch S and thus to a bridge short circuit. If DK is implemented, the Miller current is derived via DK, thus preventing a transload of C1 or a bridge short circuit.This circuit extension by DK is used for the design of the Ansteuerbrückenzweiges in open or closed form. Furthermore, it is irrelevant whether the drive circuit is supplied unipolar or bipolar.
    2. To prevent an excessive reduction of uC1 and zC2 within the turn-off interval, which would increase the risk of parasitic turn-on by a Miller current, the switch-on current path (parallel to C1) is carried out in such a way that a defined voltage is stationary at C1 UC1min remains. In the event that the switch-on current path is directly parallel to C1, this can be achieved for both the open and closed design of the control bridge branch simply by a zener diode ZD3 with a zener voltage UZ3 ≥ UC1min in series with R3 (see FIG is oriented that when flowing the Einschaltruhestroms the zener voltage causes the occurrence of a voltage of positive polarity at C1. It should be noted that when inserting the Zener diode ZD3 only the difference U1- (UZ3 + UD2) remains on the resistors in the Einschaltruhestrompfad resistors and R3 must be selected so that the required Einschaltruhestromwert is achieved. For closed execution of the Ansteuerbrückenzweiges the Einschaltruhestromwiderstand R3 with series Zener diode ZD3 can also be arranged directly branching from the output A of the Ansteuerbrückenzweiges without the above-described function is impaired.Alternatively, when the drive bridge branch is open, deep discharge of C1 can also be prevented by inserting a diode DB1 in the current flow direction against C1 and placing the turn on current resistor R3 not directly parallel to C1 but from the node between R1 and DB1 against the upper control terminal 1 is (see Fig. 3), or between S1 and R1. Thus, within the turn-off interval, a discharge of C1 across R1 and R3 is blocked by DB1. Thus, no zener diode ZD3 is required, which is advantageous in particular for small supply voltages U1 and thus capacitor voltage values uC1.
    3. Arrangement of a voltage limiting branch parallel to the control input of the transistor, wherein a physically directed from the lower control terminal 2 against the upper control terminal 1 voltage to a maximum value U21lim is limited (see Fig. 3 and Fig. 4). After applying a turn-off command, C1 is then rapidly discharged from U1-UD2 to U21lim and, on the one hand, ensures a sufficiently high negative gate-source voltage and, on the other hand, a sufficiently high difference of uC1 with respect to U1, which at the subsequent switch-on initiates an initially high inrush current via S1, R1, C1, R2, C2 drives and thus guarantees a fast charging of C2 to uC2 = Uth. In the simplest case, the voltage limiting branch can be realized by a Zener diode ZD12 with serial diode SD12, wherein the Zener voltage and also the series diode in the current flow direction are physically directed from the lower control connection 2 to the upper control connection 1. For zener voltage UZD12 of ZD12, a value U21lim-UFSD is then to be selected, where UFSD designates the forward voltage of SD12. Advantageously, a high capacitance value can thus be selected for C1, with which a change of uC1 by a Miller current within the switch-off interval can be kept low. Without the voltage limiting branch, the discharge of UC1 would be relatively slow after the initial voltage decrease as a result of the reloading of C2, as determined by the values of R4 and R2 (when the bridge branch arm is in open bridge branching), and thus with a short turn-off time. To low voltage reduction take place, so that the next time the switch S only a relatively low initial charging current for C2 would be available.
    4. Establishing a defined initial voltage at C1 prior to the first switch-on of the switch S (the capacitor C1 is brought to a voltage uC1> 0 before the first switch-on): a transistor T12 (not shown in the figures) or in general a connection switch is parallel to the control input, ie arranged in the current flow direction between the upper control input 1 and lower control input 2. The connection switch is turned on by S1 the first time, so that C1 can be charged to a positive voltage without the switch S being turned on. Thus, a negative voltage uC2 is then produced when switching off, which prevents a turn on due to the Millereffekts. The transistor T12 can advantageously be placed anti-parallel to SD12 (see FIG. 3), whereby an explicit diode SD12 u. can be omitted, since in the implementation of T12 by a MOSFET, this has a parasitic antiparallel internal diode. The transistor T12 can be switched off after an on-off game or only after several on-off games or generally when a sufficiently high voltage is established at C1.
    A modification of the drive circuit according to FIG. 4, which ensures a negative internal control voltage uC2 even before the first switching on of the electronic switch and thus reliably keeps the electronic switch in the switched-off state, is shown in FIG. Furthermore, there are characteristic time profiles of the control signals S1 and S2 of the transistors of the control bridge branch and the voltage uC1 of the coupling capacitor C1 and the internal drive voltage uC2 of the electronic switch S to be controlled.
    The parallel circuit of coupling capacitor C1 and clamping diode DK and the series circuit of Zener diode ZD3 and resistor R3 (hereinafter also referred to as coupling unit) is thereby from the connection of the output of the Ansteuerbrückenzweiges S1 and S2 with the upper control terminal 1 in the compound of the lower Control terminal 2 shifted with the negative terminal of the drive supply voltage source U1, in which case the positive terminal of the coupling capacitor C1 and the cathode of the clamping diode DK and Zener diode ZD3 come to rest on the lower control terminal 2; Further, a charging resistor R5 is inserted between the positive terminal p of the driving supply voltage source U1 and the lower control terminal 2.
    If the coupling unit in the switched path, this is referred to as the first circuit variant, the coupling unit is in the non-switched path to the negative terminal, this is referred to as the second circuit variant.
    With the occurring at the beginning of the startup of the drive circuit applying the Ansteuerversorgungsspannung U1 in a terminal time t-1 then the coupling capacitor C1 is charged via the charging resistor R5 to a positive, defined by ZD3 voltage UZD3, the transistors S1 and S2 are located in switched off state. With an applied drive supply voltage U1, a local signal processing unit, not shown in FIG. 5, then begins to operate in t0; S2 is switched through by this signal processing unit, S1 remains blocked for the time being. Thus, the voltage limiting branch formed by the series connection of Zener diode ZD21 and series diode SD21 and the parallel circuit of on-resistance R1 and off resistance R4 with series diode D4 are parallel to C1 and the voltage on C1 is set to the Zener voltage of ZD21, i. lowered to U21lim. The voltage then applied between lower control terminal 2 and upper control terminal 1 leads to a charging of the inner drive-side internal equivalent capacitance C2 of the electronic switch S to a negative control voltage uC2 = -U21lim and thus to a safe locking of the switch S until a second time t2 of FIG actual switching operation of the electronic switch S begins.
    In the second time t2 then S2 is turned off and S1 is turned on, and thus the full Ansteuerversorgungsspannung U1 to the series circuit of the inner drive side equivalent circuit of the switch S - formed from inner drive resistor R2 and the parallel circuit of internal equivalent capacitance C2 and diode D2 - and the coupling capacity C1 laid. The resulting charge of C2 on the forward voltage UD2 of D2 leads to exceeding the Einschaltschwellspannung Uth and thus to a switching on the electronic switch S. The differential voltage U1-UD2 is taken over according to the series circuit of the coupling capacitor C1, i. C1 charged to uC1 = U1-UD2.
    As an alternative to the processes described above, the startup of the drive circuit can also be such that the drive supply voltage is applied only at a start time t0 and defacto simultaneously the local signal processing unit starts to work and S2 turns on and S1 blocks. The voltage uC1 at C1 is then built up again via the charging resistor R5, but now the voltage limiting element (series connection of ZD21 and SD21) lying between the negative control terminal 2 and the positive control terminal 1 determines the voltage end value, ie only reaches a final voltage value U21lim in a first instant t1 becomes. The processes from the second time t2 then correspond to the ratios described above, as shown in FIG.
    For the switching off of the electronic switch S1 is blocked in a third time t3 and S2 turned on, whereby the voltage U1-UD2 initially at C1 in the direction of a switch blocking the drive voltage ust to the series circuit - formed from the inner control-side equivalent circuit of the switch S with parallel limiting branch ZD21 and SD21 and the parallel connection of R1 and R4 with series diode D4 - is placed. Accordingly, C2 is reloaded to a negative voltage U21lim, i. that a negative, in any case below the switch-on threshold Uth lying inner control voltage uC2 = -U21lim occurs and the electronic switch S is securely locked. As described above, the voltage U21lim then also occurs at C1, but in the positive direction uC1 = U21lim, which again means that the same conditions prevail as before the first switch-on in t2.
    The function of the clamping diode DK as well as the remaining function of the series circuit of ZD3 and R3 remains as described for Fig. 4 and is therefore not further discussed here.
    5, an embodiment of the drive switching stage is also possible as an open bridge branch (compare FIG. 1) and with a coupling unit connected in the connection of the lower control terminal 2 to the negative terminal of the drive supply voltage source U1. In this case, the upper series resistance (on-resistance) R1 is then directly connected to the terminal of the upper drive transistor S1 facing away from the positive terminal p of the drive supply voltage source U1 and the diode D4 can be omitted, and the lower series resistance R4 (turn-off resistance) is applied directly to the negative Terminal n the drive supply voltage remote terminal of the lower drive transistor S2 set.
    Remarks:
    [0040]<tb> • <SEP> The main elements 1.), 2.), 3.) are separately usable separately and in any combination. Likewise, they can be used analogously in the first as well as the second circuit variant.<tb> • <SEP> The serial diode for preventing the discharge of C1 in the switch-off interval can only be used with an open control bridge branch structure. Alternatively, a Zener diode ZD3 can be used in series with R3; when the structure of the Ansteuerbrückenzweiges closed only the Zener diode ZD3 is useful.<tb> • <SEP> 4.) is used to establish a defined initial state, thus describing a measure which is no longer effective in stationary operation.
    权利要求:
    Claims (5)
    [1]
    1. A drive circuit for driving a power transistor, wherein the drive circuit<tb> • <SEP> has an upper terminal (p) and a lower terminal (0) of a supply terminal for connection to a unipolar drive supply voltage source,<tb> • <SEP> has a first control terminal (1) and a second control terminal (2) for connection to an electronic switch to be controlled;<tb> • <SEP> has a coupling capacitor (C1) and a switching stage with electronic switches (S1, S2), wherein<tb> • <SEP> in a first circuit variant by means of the switching stage, a first current path from the first control terminal (1) through the coupling capacitor (C1) is selectively formed to the upper terminal (p) or lower terminal (0)<tb> <SEP> - <SEP> and the drive circuit has a switch-on current resistor (R3) which is connected in a second current path from the first control connection (1) to the same terminal of the supply connection to which the first current path leads, and parallel to the coupling capacitor ( C1) is arranged,<tb> • <SEP> in a second circuit variant by means of the switching stage, a first current path from the first control terminal (1) optionally to the upper terminal (p) or the lower terminal (0) can be formed and the coupling capacitor (C1) between the second control terminal ( 2) and the lower terminal (0) is connected,<tb> <SEP> - <SEP> and the drive circuit has a switch-on current resistor (R3), which is arranged in a second current path from the second control connection (2) to the lower terminal (0) of the supply connection, and parallel to the coupling capacitor (C1),<tb> wherein both circuit variants are characterized in that at least one of the following elements is present:<tb> • <SEP> a limiting element which limits a voltage across the coupling capacitor (C1) to values of only one polarity;<tb> • <SEP> a discharge limiting element which limits discharge of the coupling capacitor (C1) by the turn-on rush current resistance (R3) to a minimum voltage;<tb> • <SEP> a control voltage limiting element which limits a voltage between the first control terminal (1) and the second control terminal (2), measured from the second to the first control terminal, to a maximum value (U21lim);<tb> • <SEP> a connection switch between the first control terminal (1) and the second control terminal (2), with an auxiliary control which is adapted to turn on the connection switch before first turning on the power transistor to charge the coupling capacitor (C1), without turning on the power transistor.
    [2]
    2. Control circuit according to claim 1, wherein the limiting element is a diode (DK), in particular a Schottky diode.
    [3]
    3. Control circuit according to claim 1 or 2, wherein the discharge limiting element is a diode which blocks a discharge of the coupling capacitor (C1) by the Einschaltruhestromwiderstand (R3).
    [4]
    4. Control circuit according to claim 1 or 2 or 3, wherein the control voltage limiting element is a series circuit of a Zener diode (ZD12) and another diode (SD12).
    [5]
    5. Control circuit according to claim 1 or 2 or 3 or 4, wherein the additional control is adapted to turn off the connection switch when a sufficiently high voltage at the coupling capacitor (C1) is constructed.
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    同族专利:
    公开号 | 公开日
    CH711457B1|2019-09-13|
    CH711455A2|2017-02-28|
    引用文献:
    公开号 | 申请日 | 公开日 | 申请人 | 专利标题
    法律状态:
    2018-11-15| PCAR| Change of the address of the representative|Free format text: NEW ADDRESS: POSTFACH, 8032 ZUERICH (CH) |
    2020-08-31| PL| Patent ceased|
    优先权:
    申请号 | 申请日 | 专利标题
    CH01240/15A|CH711455A2|2015-08-28|2015-08-28|Device for the pulse-duration-independent, reliable control of power transistors.|
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