Method and control system of a brushless permanent magnet motor, and motor system

专利摘要:
control of a brushless permanent magnet motor. It is a method of controlling a brushless permanent magnet motor, the method comprising rectifying an alternating voltage to provide a rectified voltage with a ripple of at least 50% and exciting a motor winding with the rectified voltage. the winding is excited before the counter electromotive force passes through zero for an advance period and is excited for a driving period during each electric motor half-cycle. The lead time and / or driving period are then adjusted in response to changes in engine speed and / or the rms value of the alternating voltage to keep the average power constant. In addition, a control system implementing the method and a motor system incorporating the control system are disclosed.
公开号:BR112012026387B1
申请号:R112012026387-0
申请日:2011-04-13
公开日:2019-11-12
发明作者:Clothier Andrew；Greetham Stephen
申请人:Dyson Technology Ltd；
IPC主号:

专利说明:

“METHOD AND CONTROL SYSTEM OF A PERMANENT MAGNET MOTOR WITHOUT BRUSHES AND ENGINE SYSTEM”
The present invention relates to the control of a brushless permanent magnet motor.
A brushless permanent magnet motor typically includes a control system to control the excitation of the motor's phase windings. When driven by AC power, the control system usually includes an active power factor correction (PEC) circuit, which generates a regular DC voltage for use in the excitation of the phase windings. With the provision of a regular DC voltage, the motor power can be controlled relatively satisfactorily. However, an active PFC circuit considerably increases the cost of the control system. In addition, the PFC circuit requires a high impedance DC link capacitor, which is both physically large and expensive, in order to provide a regular feedback voltage to the PFC circuit.
In a first aspect, the present invention proposes a method of controlling a brushless permanent magnet motor, the method comprising: rectifying an alternating voltage to provide a rectified voltage with a ripple of at least 50%; excite a motor winding with the voltage rectified, the winding being excited before zero crossings of the electromotive force for a period of advance and being excited for a driving period during each electric motor semicycle; and adjusting one of the lead time and driving time in response to changes in one of the motor speed and the RMS value of the AC voltage in order to keep the average power constant.
By adjusting one of the advance period and driving period, or both, it is possible to achieve a constant average power despite the ripple in the alternating voltage. Consequently, the constant average power can be achieved without the need for an active PFC or a high capacitance bonding capacitor.
The lead time and / or lead time can be adjusted so that one or both of the average input and average output powers are kept constant. It should be understood that “constant average power”, in this case, means that the variation in average power is not greater than +/- 5%.
Since the motor is driven by an alternating voltage, the instantaneous power will vary during each cycle of the alternating voltage. Therefore, the reference to the average power must be understood as the motor power (input or output power) whose average was calculated during an alternating voltage cycle.
The lead time and / or lead time can be adjusted so that an efficiency (i.e., the ratio of the output power to the input power) of at least 80% is maintained. Therefore, it is possible to create a high efficiency motor, with
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2/57 constant.
The alternating voltage can be supplied by a mains supply, for which the RMS voltage is between 100 V and 240 V. The advance period and / or the driving period are then adjusted in order to maintain a constant average power of at least 1000 W. Therefore, it is possible to obtain a relatively high power motor using a mains supply.
The lead time and / or driving time can be adjusted so that a constant average power is maintained over a speed range covering 10 krpm and / or a voltage range covering 10 V. Therefore, the constant average power is maintained over a relatively wide range of motor speeds and / or RMS voltages.
The speed range can have a minimum krpm greater than 60 and a maximum krpm greater than 80. Furthermore, the maximum can be greater than 100 krpm. Therefore, the constant average output power can be obtained at relatively high engine speeds.
The length of the lead-in period and / or the driving period can be defined by a waveform that varies over each alternating voltage semicycle. Each of the length of the advance period and driving influence the amount of current transmitted to the winding during an electric motor cycle, which in turn influences the amount of current drawn from the power supply that supplies the alternating voltage . The waveform of the advance period and / or the driving period can therefore be defined so as to conform the waveform of the current drawn from the power supply. In particular, the waveform of the advance period and / or the driving period can be defined so that the waveform of the current drawn from the power supply is close to that of a sinusoid. Consequently, a relatively higher power factor can be obtained without the need for a PFC circuit or a high capacitance bonding capacitor. The waveform of the lead time and / or driving time is then adjusted in response to changes in engine speed and / or RMS voltage. Consequently, a constant power motor with a relatively high power factor can be realized.
The method may comprise determining the lead time and / or driving period for each electric motor semicycle. This then helps to obtain a more uniform envelope for the current waveform drawn from the power supply. It is not necessary that a different lead time and / or driving time be used for successive electric motor cycles. Depending on the shape of the waveform for the advance period and / or the driving period, as well as the duration of each engine electric semicycle, it is reasonably possible that the successive electric semicycle of the engine have the same advance period and / or driving period. For example, if the
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3/57 waveform of the advance period and / or the driving period included a flat point, so it would be quite possible that two successive electric motor cycles had the same advance period and / or driving period.
The variation in the waveform of the conduction period can take the form of a triangle, trapezoid or semi-sinusoid during each cycle of the waveform. The driving period then generally increases during the first half of the cycle and decreases during the second half of each half cycle of alternating voltage. In contrast, the variation in the waveform of the advance period can take the form of an inverted triangle, an inverted trapezoid or an inverted semi-sinusoid during each waveform cycle. The lead time then generally decreases during the first half of the cycle and increases during the second half of each half cycle of alternating voltage. It was found that each of these waveforms is effective in obtaining a current waveform that approximates that of a sine wave, thus resulting in a relatively high power factor.
The length of the lead time and / or driving time can be defined as the sum of a first component and a second component. The first component is then constant during each AC cycle, and the second component varies during each AC cycle. The first component, in this way, acts as a deviation or elevation for the waveform of the advance period or the driving period. Consequently, a higher average input power can be achieved for a given peak current. The method then comprises adjusting the first component in response to changes in motor speed and / or RMS voltage. Defining the lead time and / or driving time as the sum of the two components is a convenient method for adjusting the waveform in response to changes in motor speed and / or RMS voltage.
The second component can be defined by the length of time that has elapsed since a zero crossing at the alternating voltage. This then ensures that the lead time and / or lead time waveform is synchronized with the voltage waveform of the alternating voltage. As a result, the current waveform drawn from the power supply is generally more stable.
The conduction period waveform may have a phase shift from the alternating voltage voltage waveform. This phase shift then acts to dampen the lower order harmonics within the current waveform. Thanks to the electromotive force induced in the winding by the rotor, the magnitude of the lower order harmonics can vary with changes in the motor speed and / or in the RMS voltage. Thus, the method may include adjusting the phase shift in response to changes in motor speed and / or RMS voltage in order to maintain harmonics
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4/57 lower order chain relatively small.
Adjusting the lead time and / or driving time (for example, in response to changes in engine speed and / or RMS voltage) at different points within each AC cycle can potentially increase the harmonic content of the current wave drawn from the power supply. In addition, when the motor is operating at a constant average speed, the instantaneous motor speed, however, varies during each half cycle of the AC voltage thanks to the sinusoidal increase and the decrease in the rectified voltage. If the lead time and / or driving period were adjusted for changes in engine speed, and the adjustment occurred at different points within each AC cycle, the lead period and / or driving period could be adjusted despite the fact that the average engine speed has not changed. Again, this can result in increased harmonics within the current waveform. The lead time and / or lead time can therefore be adjusted in response to a zero crossing of the AC voltage. As a result, the same reference point in the AC cycle is used. Consequently, it is possible to obtain a more stable current waveform. Furthermore, by adjusting the advance period and / or the driving period no more than once for each alternating voltage semicycle, the motor control is kept relatively simple, and thus a relatively simple and Cheap can be used to implement the method.
The lead time can be constant during each alternating voltage semicycle and the method can comprise adjusting the lead time in response to a zero crossing at the alternating voltage. The same advance period is then used for each electric motor semicycle encompassing an alternating voltage semicycle. This then further simplifies engine control.
The lead time and / or lead time can be stored as one or more look-up tables. For example, the method may comprise storing a first lookup table of first control values, which is then indexed in response to a zero crossing at the alternating voltage. The lookup table is indexed using one of the speed and voltage to select a first control value, which is then used to determine the lead time. The duration of the driving period can comprise the sum of the first component that is constant and a second component that varies during each alternating voltage semicycle. The method can then comprise storing a second lookup table for second control values, which is then indexed in response to a zero crossing at the alternating voltage. The second lookup table is indexed using one of the speed and voltage to select a second control value, which is then used to determine the first component. O
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5/57 use of look-up tables to determine the advance period and / or the driving period simplifies engine control.
The control values stored by the lookup tables can be absolute values or difference values. When the control values are difference values, the method also comprises storing a reference value to which the difference value is applied to obtain the lead time and / or the lead time. Storing difference values usually requires less memory than absolute values, so query tables can be stored more efficiently.
In a second aspect, the present invention proposes a control system for a brushless permanent magnet motor, the control system carrying out the method as described in any of the previous paragraphs.
The control system can comprise a rectifier to rectify the alternating voltage, an inverter coupled to the winding and a controller to control the inverter. The controller then generates one or more control signals to excite the winding before zero crossings of the electromotive force. The inverter, in response to the control signals, then excites the winding with the rectified voltage. The controller then adjusts the lead time and / or driving time in response to changes in engine speed and / or RMS voltage in order to keep the average power constant.
The control system may include a position sensor that emits a signal having edges that correspond to zero crossings of the force against the electromotive in the winding. The controller then generates the control signals before each edge of the signal.
The control system may comprise a zero crossing detector to detect zero crossings at the alternating voltage. The controller then adjusts the lead time and / or lead time in response to zero crossings.
In a third aspect, the present invention proposes a motor system comprising a brushless permanent magnet motor and a control system as described in any of the previous paragraphs.
In order that the present invention can be more easily understood, embodiments of the invention will now be described, by way of example, with reference to the accompanying drawings, in which:
Figure 1 is a block diagram of an engine system according to the present invention;
Figure 2 is a schematic diagram of the engine system;
Figure 3 is a sectional view of the engine of the engine system;
Figure 4 details the allowed states of the inverter in response to the control signals emitted by the motor system controller;
Figure 5 is a schematic diagram of the current regulator of the handheld
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Figure 6 illustrates the overheating period used by the controller when operating in single switching mode;
Figure 7 illustrates a three-step process used by the controller when measuring analog input signals;
Figure 8 details the various modes of operation of the engine system;
Figure 9 details the direction in which the motor is started in response to the control signals emitted by the controller;
Figure 10 illustrates various engine system waveforms while operating in Low Speed Acceleration Mode;
Figure 11 illustrates various waveforms of the engine system while operating in High Speed Acceleration Mode;
Figure 12 illustrates various waveforms of the engine system while operating in Travel Mode;
Figure 13 illustrates the current waveform drawn from the motor system power supply while operating in Travel Mode;
Figure 14 illustrates various waveforms and interruptions of the motor system while operating in single overcurrent switching mode;
Figure 15 illustrates various waveforms and interruptions of the motor system while operating in single unlimited free-spin switching mode;
Figure 16 is a schematic diagram of a timer and a comparator module configured to generate a control signal;
Figure 17 is a schematic diagram of a timer and a PWM module configured to generate a control signal;
Figure 18 illustrates various waveforms and interruptions of the engine system while operating in single, unlimited free-spin switching mode;
Figure 19 details the values of the various hardware components for a particular embodiment of the engine system according to the present invention;
Figure 20 details several constants and thresholds employed by the controller of the particular engine system;
Figure 21 illustrates the flow connection characteristics of a motor system connection inductor in particular;
Figure 22 illustrates the flow connection characteristics of the motor system inductor in particular;
Figure 23 details the various modes of operation of the particular engine system;
Figure 24 details a map of the control values used by the controller of the
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7/57 private engine system while operating in multi-switching mode;
Figure 25 details a map of the control values used by the particular motor system controller while operating in single overcurrent switching mode;
Figure 26 details a part of the forward lookup table used by the controller of the particular engine system while operating in the single unlimited free-spin switching mode;
Figure 27 details a portion of the bypass lookup table used by the controller of the particular engine system while operating in the unlimited free-spin single switching mode;
Figure 28 details a part of the phase look-up table used by the controller of the particular motor system while operating in the unlimited free-spin single switching mode;
Figure 29 details a portion of the sinusoidal map used by the motor system controller in particular while operating in single switching mode;
Figure 30 illustrates possible waveforms for the driving period used by the controller in single switching mode; and
Figure 31 illustrates possible waveforms for the advance period of an alternative engine system according to the present invention.
The motor system 1 of Figures 1 to 3 comprises a brushless motor 2 and a control system 3. The power for motor system 1 is powered by AC power 4. AC power 4 is assumed to be a network domestic power supplies, although other power supplies capable of supplying an alternating voltage can also be used.
Motor 2 comprises a tetrapolar permanent magnet rotor 5 that rotates in relation to a stator 6. Stator 6 comprises a pair of “C” shaped cores that define four poles of the stator. The conductor wires are wound around the stator 6 and are coupled together to form a single phase winding 7.
The control system 3 comprises a rectifier 8, a DC link filter 9, an inverter 10, a door trigger module 11, a current sensor 12, a position sensor 13, a zero crossing detector 14, a sensor temperature controller 15 and a controller 16.
Rectifier 8 is a full wave bridge D1-D4 that rectifies the output of AC power 4 to provide a DC voltage.
The DC link filter 9 comprises a link capacitor C1 and a link inductor L1. The connection capacitor C1 acts to smooth out the relatively high frequency ripple that comes from switching the inverter. As described in more detail below, the C1 bonding capacitor does not need to smooth the DC voltage rectified at the
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8/57 fundamental company. Consequently, a relatively low capacitance bonding capacitor can be used. The connection inductor L1 acts to smooth out any residual current ripple that comes from switching the inverter. Again, since the link inductor L1 is intended to reduce the ripple in the switching frequency of the inverter 10, a relatively low inductance inductor can be used. So that saturation is avoided, the link inductor L1 has a saturation point that exceeds the peak current drawn from AC power 4 during normal operation of motor system 1.
The inverter 10 comprises a complete bridge of four power switches Q1-Q4 that couple the DC link voltage to the phase winding 7. Each power switch Q1-Q4 is an IGBT (Bipolar Transistor Isolated), which is capable of operate at the voltage level typical of most mains power supplies. Other types of power switches, such as BJTs or MOSFETs, could be used alternatively, depending on the rated power of the power switch and the voltage of the AC power 4. Each of the Q1-Q4 switches includes a return diode, which protects the switch against voltage spikes that arise during switching of the inverter.
The door trigger module 11 triggers the opening and closing of the Q1Q4 switches of the inverter 10 in response to the control signals received from the controller 16.
Current sensor 12 comprises a pair of shunt resistors R1, R2, each resistor located on a lower arm of the inverter 10. The resistance of each shunt resistor R1, R2 is preferably the highest possible, without exceeding the limits dissipation during normal motor system operation 1. The voltage across each shunt resistor R1, R2 is output to control 16 as a current detection signal, I_SENSE_1 and I_SENSE_2. The first current detection signal, I_SENSE_1, provides a measurement of the current in the phase 7 winding when driven from right to left (as described in more detail below). The second current detection signal, I_SENSE_2, provides a measurement of the current in the phase 7 winding when driven from left to right. When locating the branch resistors R1, R2 in the lower arms of the inverter 10, the current in the winding in phase 7 continues to be detected during the free wheel (again, as described in more detail below).
Position sensor 13 is a Hall effect sensor that emits a digital signal, HALL, which is logically high or low, depending on the direction of the magnetic flux through sensor 13. When locating position sensor 13 adjacent to rotor 5, the HALL signal provides a measurement of the angular position of rotor 5. More particularly, each edge of the HALL signal indicates a change in the polarity of rotor 5. When rotating, the permanent magnet rotor induces a force against the electromotive in the phase 7 winding. each edge of the HALL signal also represents a change in the polarity of the force against the electromotive in the phase 7 winding.
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The zero crossing detector 14 comprises a pair of fixing diodes D5, D6 that emit a digital signal, Z_CROSS, which is logically high when the AC supply voltage is positive, and logically low when the AC supply voltage is negative . Each edge of the Z_CROSS signal thus represents a point in time when the voltage of the AC supply 4 passes through zero.
The temperature sensor 15 comprises a thermistor R7 that emits an analog signal, TEMP, which provides a temperature measurement within the motor system 1.
Controller 16 comprises a microcontroller having a processor 17, a memory device 18, a plurality of peripherals 19 (e.g., ADC, comparators, timers, etc.), a plurality of input pins 20 and a plurality of output pins 21. The memory device 18 stores software instructions for execution by the processor 17. The memory device 18 also stores a plurality of look-up tables, which are indexed by the processor 17 during the operation of the engine system 1.
Controller 16 is responsible for controlling the operation of motor system 1. In response to signals at input pins 20, controller 16 generates control signals at output pins 21. Output pins 21 are coupled to the door actuator module 11, which controls the opening and closing of the Q1-Q4 switches of the inverter 10 and the response to the control signals.
Seven signals are received at input pins 20 of controller 16: I_SENSE_1, I_SENSE_2, HALL, Z_CROSS, TEMP, DC_LINK and DC_SMOOTH. I_SENSE_1 and I_SENSE_2 are the signals emitted by the current sensor 12. HALL is the signal emitted by the position sensor 13. Z_CROSS is the signal emitted by the zero crossing detector 14. TEMP is the signal emitted by the temperature sensor 15. DC_LINK is a proportionally reduced measure of the DC link voltage, which is obtained by a potential divider R3, R4 located between the DC link line and the zero volt line. DC_SMOOTH is a smoothed measurement of the DC link voltage, obtained by a potential divider R5, R6 and the smoothing capacitor C2.
In response to the signals received at the inputs, controller 16 generates and emits four control signals: TRIP #, DIR1, DIR2, and FREEWHEEL #.
TRIP # is a safety control signal. When TRIP # is pulled logically low, the door actuator module 11 opens all the switches Q1-Q4 of the inverter 10. As described in more detail below, controller 16 pulls TRIP # logically low in case the current through the phase winding 7 exceeds a safety threshold.
DIR1 and DIR2 control the direction of the current through the inverter 10, and therefore through the phase winding 7. When DIR1 is pulled logically high and DIR2 is pulled logically low, the door actuator module 11 closes the switches Q1 and Q4 opens at
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10/57 switches Q2 and Q3, thus causing the current to be activated through the winding in phase 7 from left to right. Conversely, when DIR2 is pulled logically high and DIR1 is pulled logically low, the door trigger module 11 closes switches Q2 and Q3 opens switches Q1 and Q4, thus causing the current to be driven through the phase 7 winding on the right to the left. The current in the winding in phase 7 is therefore switched by reversing DIR1 and DIR2. If both DIR1 and DIR2 are pulled logically low, door trigger module 11 opens all switches Q1-Q4.
FREEWHEEL # is used to disconnect the phase 7 winding from the DC link voltage and allow the current in the phase 7 winding to recirculate or rotate freely around the closed circuit on the low side of the inverter 10. Therefore, in response to a FREEWHEEL signal # which is pulled logically low, the door trigger module 11 causes both switches Q1, Q2 on the high side to open. The current then spins freely around the low-side closed loop of the inverter 10 in a direction defined by DIR1 and DIR2.
Figure 4 summarizes the permitted states of switches Q1-Q4 in response to control signals from controller 16. Hereafter, the terms “define” and “release” will be used to indicate that a signal has been pulled logically high and low, respectively .
When a particular control signal changes, there is a short delay between changing the control signal and the physical opening or closing of a power switch. If an additional control signal were changed during this delay period, it is possible that both switches on a particular drive arm (ie Q1, Q3 or Q2, Q4) could be closed at the same time. This short, or “trip” as it is generally called, would damage the switches Q1, Q2 in this arm of the inverter 10. Therefore, in order to avoid the trip, the controller 16 uses an idle time, T_DT, between the alteration of the two control. Then, for example, during switching the winding in phase 7, controller 16 first releases DIR1, waits for idle time, T_DT, and then sets DIR2. Idle time is preferably kept as short as possible in order to optimize performance, while also ensuring that door trigger module 11 and power switches Q1-Q4 have sufficient time to respond.
SWITCHING
Controller 16 switches the winding in phase 7 in response to the edges of the HALL signal. Switching involves reversing DIR1 and DIR2 (ie, releasing DIR1 and defining DIR2, or releasing DIR2 and defining DIR1) in order to reverse the direction of the current through the phase 7 winding. winding in phase 7 may be rotating free-wheel at the switching point. Therefore, in addition to the inversion of DIR1 and DIR2, controller 16 defines FREEWHEEL #.
Synchronous switching
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Below a predetermined speed threshold, SPEED_ADV, controller 16 switches the winding in phase 7 in sync with the edges of the HALL signal. Each edge of the HALL signal also represents a change in the polarity of the force against the electromotive in the phase 7 winding. Consequently, at speeds below SPEED_ADV, controller 16 switches the winding in phase 7 in sync with the zero crossings of the electromotive force.
As the rotor 5 accelerates, the period of each electric semicycle decreases, and, thus, the time constant (L / R) associated with the inductance of the winding in phase 7 becomes increasingly important. In addition, the magnitude of the counter electromotive force in the phase 7 winding increases, which then influences the rate at which the current increases in the phase 7 winding. Consequently, if controller 16 continues to switch the phase 7 winding in sync with the edges of the HALL signal, a speed would be reached at which it would no longer be possible to conduct additional current to the phase 7 winding during each electric semicycle. Therefore, upon reaching SPEED_ADV, controller 16 switches from synchronous switching to advanced switching. When switching the winding in phase 7 before the edges of the HALL signal, the voltage used to excite the winding in phase 7 is intensified by the force against the electromotive. As a result, the direction of the current through the phase 7 winding can be reversed more quickly. In addition, the phase current can be induced to conduct the force against the electromotive, which then helps to compensate for the slower rate of current increase. Although this then generates a short period of negative torque, this is usually more than offset by the consequent gain in positive torque.
Advanced switching
At speeds at the speed threshold or above, SPEED_ADV, controller 16 switches the winding in phase 7 before each edge of the HALL signal for an advance period, T_ADV. Since the period of the electric semicycle decreases and the force against the electromotive increases with the speed of the rotor, the electrical angle at which the switching takes place before the edges of the HALL signal preferably increases with the speed of the rotor. For a particular lead time, T_ADV, the corresponding lead angle, A_ADV, can be defined as:
A_ADV (degrees el.) = T_ADV (sec) * {ω (rpm) / 60} * 360 (degrees mec.) * N / 2 where A_ADV is the lead angle in electrical degrees, T_ADV is the lead period in seconds, ω is the rotor speed in rpm, and n is the number of rotor poles. From this equation, it can be seen that the forward angle is directly proportional to the rotor speed. Consequently, even for a fixed feed period, the feed angle increases with the speed of the rotor. However, it is possible to obtain better control over acceleration, power and efficiency using different lead times at
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12/57 different rotor speeds. Controller 16, therefore, comprises an advance lookup table that stores an advance period for each of a plurality of rotor speeds.
In response to an edge of the Z_CROSS signal, controller 16 selects, from the advance lookup table, an advance period, T_ADV, corresponding to the speed of rotor 5. The speed of rotor 5 is determined from the interval, T_HALL , between two successive edges of the HALL signal. This interval will henceforth be called the Hall period. The speed of rotor 5 is then defined by:
ω (rpm) = 60 / {n * T_HALL (sec)} where ω is the rotor speed in rpm, T_HALL is the Hall period in seconds, and n is the number of rotor poles. Controller 16 uses the selected lead time to switch the winding in phase 7 before the edges of the HALL signal. The same lead time, T_ADV, is then used by controller 16 until such time that an additional edge of the Z_CROSS signal is detected. In response to an additional edge of the Z_CROSS signal, controller 16 selects, from the advance lookup table, a new advance period corresponding to rotor speed 5. The advance period is therefore updated only when the voltage of the AC 4 supply goes through zero, and is constant during each AC 4 semiconductor cycle.
In order to switch the winding in phase 7 before a particular edge of the HALL signal, controller 16 acts in response to the leading edge of the HALL signal. In response to an edge of the HALL signal, controller 16 subtracts the lead period, T_ADV, from the Hall period, T_HALL, in order to obtain a switching period, T_COM:
T_COM = T_HALL - T_ADV
Controller 16 then switches the winding in phase 7 at a time, T_COM, after the edge of the HALL signal. As a result, the phase 7 winding is switched before the subsequent edge of the HALL signal by the lead time, T_ADV.
As noted above, the advance period, T_ADV, remains fixed during each AC 4 power cycle. However, rotor speed 5 varies over each AC 4 power cycle thanks to the sinusoidal increase and the decrease in the connection voltage. CC. The Hall period, T_HALL, therefore, varies during each AC power semiconductor 4. Consequently, different from the advance period, controller 16 calculates the switching period, T_COM, for each edge of the HALL signal.
CURRENT CONTROL
Several of the peripherals 19 of controller 16 are configured to define a current regulator 22. Current regulator 22 monitors and regulates the current in the phase 7 winding. Current regulator 22 performs two functions. First, the current regulator 22 releases TRP # in the event that the current in the winding in phase 7 exPetition 870190080086, from 19/08/2019, pg. 19/93
13/57 yields a security threshold. Second, current regulator 22 generates an overcurrent signal in the event that the current in the phase 7 winding exceeds an overcurrent threshold.
As illustrated in Figure 5, the current regulator 22 comprises a safety module 23 and an overcurrent module 24.
The security module 23 comprises a multiplexer 25, a comparator 26, a NOT port 27 and an SR 28 coupling. Multiplexer 25 has two inputs for selecting one of the two current detection signals, I_SENSE_1 and I_SENSE_2. The selection made by multiplexer 25 is controlled by processor 17 in response to the current direction through the phase 7 winding. In particular, when DIR1 is defined, multiplexer 25 is prompted to select I_SENSE_1, and when DIR2 is defined, the multiplexer is prompted to select I_SENSE_2. The output of multiplexer 25 is distributed to comparator 26, which compares the voltage of the selected current detection signal with a predetermined safety voltage, TRIP_REF. TRIP_REF is defined so that the output of comparator 26 is pulled logically high when the current through the selected shunt resistor R1, R2 is greater than a predetermined safety threshold I_MAX. TRIP_REF is therefore defined by I_MAX and the resistances of the resistors in derivation R1, R2. The output of comparator 26 is distributed to the NOT 27 port, the output of which is distributed to input S of the SR 28 coupling. The Q # output of the SR 28 coupling is output by the current regulator 22 as the TRIP # signal. Consequently, when the current detection signal voltage, I_SENSE_1 or I_SENSE_2, is greater than TRIP_REF, TRIP # is released.
As noted above, the door trigger module 11 opens all the Q1Q4 switches of the drive 10 in response to a released TRIP # signal. The safety module 23 of the current regulator 22 therefore prevents the current in the winding in phase 7 from exceeding a safety threshold, I_MAX, above which the Q1-Q4 switches can be damaged and / or the rotor 5 can be demagnetized . By using hardware to release the TRP # signal, the current regulator 22 responds relatively quickly when the current in the phase 7 winding exceeds the safety threshold. If the software executed by processor 17 were used instead to release the TRIP # signal, a delay could occur between the current exceeding the safety threshold and the release of the TRIP # signal, during which time the current can rise to a level that damages keys Q1-Q4 or demagnetizes rotor 5.
Processor 17 interrogates the TRIP signal # in response to each edge of the HALL signal. If the TRIP # signal is released for five successive HALL edges, processor 17 writes a “Safety Margin Exceeded” error to memory device 18 and enters Fault Mode, which is described in more detail below. Monitoring the TRIP # signal in this way ensures that controller 16 does not accidentally enter Failure Mode due to
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The overcurrent module 24 comprises a multiplexer 29 and a comparator 30. Multiplexer 29, like that of the safety module 23, has two inputs for selecting one of the two current detection signals, I_SENSE_1 and I_SENSE_2. Again, the selection made by multiplexer 29 is controlled by processor 17 in response to the current direction through the phase 7 winding. Consequently, when DIR1 is defined, multiplexer 29 selects I_SENSE_1, and when DIR2 is defined, multiplexer 29 selects I_SENSE_2 . The output from multiplexer 29 is distributed to comparator 30, which compares the voltage of the current detection signal with the voltage of the DC_LINK signal. When the current detection signal, I_SENSE_1 or I_SENSE_2, is greater than DC_LINK, comparator output 30 is pulled logically low. The overcurrent module 24 thus emits an overcurrent signal that is pulled logically low when the current in the phase 7 winding exceeds an overcurrent threshold that is proportional to the DC link voltage.
The output of the overcurrent module 24 is coupled to the processor 17, which performs an overcurrent routine in response to a low overcurrent signal. Since the overcurrent threshold is proportional to the DC link voltage, the overcurrent threshold varies as a rectified sine wave throughout each cycle of the AC 4 supply, the benefits of which are explained in more detail below.
The resistors of the potential divider R3, R4 are selected so that the peak voltage of the DC_LINK signal does not exceed TRIP_REF. Consequently, the current regulator 22 triggers an overcurrent event before the current in the phase 7 winding exceeds the safety threshold. Therefore, overcurrent module 24 and processor 17 are expected to regulate the current in the phase 7 winding. Safety module 23 is expected to release TRIP # only in the unlikely event of a failure within processor 17 (for example, example, a software failure) or if the current in the phase 7 winding rises at a speed such that the safety threshold, I_MAX, is reached before processor 17 is able to respond to the overcurrent event.
In response to the overcurrent event, controller 16 performs a different series of actions, depending on the speed of rotor 5. At speeds below a predetermined threshold, SPEED_SINGLE, controller 16 operates in a “multi-switching mode”. At speeds at or above the predetermined threshold, controller 16 operates in a “single switching mode”.
Multi-Switching Mode
In response to an overcurrent event in a multi-switching mode, controller 16 freely rotates the winding in phase 7 by releasing FREEWHEEL #. The free spin continues for a period of free spin, T_FW, time during
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15/57 which the current in the phase 7 winding is expected to decline to a level below the overcurrent threshold. If the current in the phase 7 winding continues to exceed the overcurrent threshold, the controller 16 once again spins the phase 7 winding freely during the free spin period, T_FW. If, on the other hand, the current in the phase 7 winding has fallen below the overcurrent threshold, controller 16 resumes the excitation of the phase 7 winding by setting FREEWHEEL #.
For a particular free spin period, T_FW, the corresponding electrical angle, A_FW, can be defined as:
A_FW (degrees el.) = T_FW (sec) * {ω (rpm) / 60} * 360 (degrees mec.) * N / 2 where A_FW is the angle of free rotation in electrical degrees, T_FW is the period of rotation free in seconds, ω is the rotor speed in rpm, and n is the number of rotor poles. Consequently, for a fixed free rotation period, the corresponding free rotation angle increases with the speed of the rotor. However, as the free rotation angle increases, the remaining period during which the current and therefore the energy is conducted into the phase 7 winding decreases. Controller 16, therefore, employs a free turning period, T_FW, which decreases with increasing rotor speed, so that the corresponding free turning angle, A_FW, does not become excessively large as rotor 5 accelerates.
Controller 16 comprises a freewheel lookup table that stores a freewheeling period for each of a plurality of rotor speeds. In response to an edge of the Z_CROSS signal, controller 16 selects, from the free spin query table, a free spin period, T_FW, corresponding to rotor speed 5. Controller 16 then uses the selected free spin period. to freely rotate the winding in phase 7 in response to an overcurrent event. The same free spin period, T_FW, is used by controller 16 until such time that an additional edge of the Z_CROSS signal is detected. In response to an additional edge of the Z_CROSS signal, controller 16 selects, from the free spin query table, a new free spin period corresponding to the speed of the rotor 5. Consequently, as well as the advance period, the Free rotation is therefore updated only when the AC 4 supply voltage goes through zero, and remains constant during each AC 4 semiconductor cycle.
Controller 16 operates in a multi-switching mode only while rotor 5 accelerates from stationary speed to SPEED_SINGLE. As such, the duration of time spent by controller 16 in multi-switching mode is relatively short. Therefore, a relatively common speed resolution can be used for the free spin lookup table without adversely affecting the power or efficiency of the motor system 1. In fact, a fixed free spin period could be perfectly used, as long as the angle
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16/57 corresponding free spin does not become excessively large as rotor 5 approaches SPEED_SINGLE.
At relatively low speeds of the rotor, the counter electromotive force induced in the phase 7 winding by the rotor 5 is relatively small. Consequently, the current in the winding in phase 7 increases to the overcurrent threshold relatively quickly. Thanks to the relatively short time it takes for the current to reach the overcurrent threshold, controller 16 will typically switch the winding in phase 7 between excitation and free spin multiple times during each electric semicycle of motor 2. This is why that controller 16 is said to operate in a multi-switching mode at speeds below SPEED_SINGLE. As the rotor speed increases, the Hall period naturally decreases. In addition, the force against the electromotive increases, and thus the time it takes for the current in the phase 7 winding to reach the overcurrent threshold increases. Consequently, the controller 16 switches the phase 7 winding between the excitation and the free spin frequently as the rotor 5 accelerates. Eventually, the speed of the rotor 5 rises to a level where the controller 16 switches the winding in phase 7 only once between the excitation and the free rotation during each electric semicycle of the motor 2.
Single Switching Mode
In response to an overcurrent event in single switching mode, controller 16 does not immediately rotate the winding in phase 7. Instead, controller 16 continues to excite the winding in phase 7 for a period of overheating, T_OVR. After the overheating period has elapsed, controller 16 freely rotates the winding in phase 7 releasing FREEWHEEL #. The free rotation then continues indefinitely until the controller 16 switches the winding in phase 7. The controller 16 thus switches the phase 7 winding from the excitation to the free rotation only once during each electric semicycle of the motor 2.
Referring now to Figure 6, the overheating period, T_OVR, is defined by the equation:
T_OVR = T_OVR_OFFSET + T_OVR_AMP * abs {sin (6)} where T_OVR_OFFSET is a deviation value, T_OVR_AMP * abs {sin (6)} is a rectified sine wave with an amplitude defined by T_OVR_AMP, and θ is the angle in the cycle AC power supply voltage 4.
The angle θ can be expressed as a time interval of a zero crossing in the AC supply voltage 4:
θ (degrees) = t (sec) * f (Hz) * 360 (degrees) where t is the time in seconds since a zero crossover on AC 4 supply, and f is the Hertz frequency of AC 4 supply. Overheating period Petition 870190080086, from 08/19/2019, p. 23/93
17/57 cement can then be defined as:
T_OVR = T_OVR_OFFSET + T_OVR_AMP * abs {sen (t * f * 360 degrees)}
To make it simple, the overheating period, T_OVR, can be considered as the sum of two components.
T_OVR = T_OVR_OFFSET + T_OVR_SINE where T_OVR_OFFSET is an overheating offset value that is independent of time, and T_OVR_SINE is an overheating sine value that is time dependent.
T_OVR_SINE is stored by controller 16 as an overheat sine lookup table. The superheat sine lookup table comprises a superheat sine value, T_OVR_SINE, for each of a plurality of times. In response to an edge of the HALL signal, controller 16 determines the time period, t, that has elapsed since the last edge of the Z_CROSS signal. Controller 16 then selects, from the superheat sine lookup table, a superheat sine value, T_OVR_SINE, corresponding to the elapsed time period. Controller 16 then adds the superheat offset value, T_OVR_OFFESET, and the superheat sine value, T_OVR_SINE, to obtain the superheat period, T_OVR.
As described in more detail below, by selecting the appropriate values for the advance period, T_ADV, the overheating deviation, T_OVR_OFFSET and the overheating amplitude, T_OVR_AMP, the efficiency of engine system 1 can be optimized for a power of average input or specific average output power. In addition, appropriate values can be selected so that the waveform of the current drawn from the AC 4 supply complies with the harmonic standards established by the regulating bodies.
Limit time
Regardless of the speed of the rotor, an overcurrent event is expected to occur at least once during each electric semicycle of motor 2. If an overcurrent event fails to occur, controller 16 would continue to excite the winding in faser 7, and, thus, the current in the phase 7 winding would continue to rise. At relatively high speeds of the rotor, the magnitude of the counter electromotive force in the phase 7 winding is relatively large. As a result, the phase 7 winding is unlikely to reach an excessive level, even in the absence of an overcurrent event. However, at relatively low speeds of the rotor, the electromotive force induced in the phase 7 winding is relatively small. As a result, the current in the phase 7 winding may rise to an excessive level in the absence of an overcurrent event. In fact, the current can rise to the safety threshold, I_MAX, which would then make the controller 16
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18/57 enter Failure Mode. Consequently, to the operator in multi-switching mode, controller 16 automatically executes the overcurrent routine after the winding in phase 7 has been constantly excited in the same direction for a period of time, T_TO. The timeout period thus acts as a safety mechanism that ensures that the maximum time period during which the phase 7 winding can be excited is limited.
As the speed of the rotor 5 increases, the magnitude of the electromotive force induced in the phase 7 winding also increases. Consequently, the rate at which the current increases in the phase 7 winding decreases with increasing speed of the rotor. Put another way, the electrical angle at which the current in the phase 7 winding rises until the overcurrent threshold increases with the speed of the rotor. Controller 16 therefore employs a timeout angle, A_TO, which increases with the speed of the rotor. For a particular timeout period, T_TO, the corresponding timeout angle, A_TO, can be defined as:
A_TO (degrees el.) = T_TO (sec) * {ω (rpm) / 60} * 360 (degrees mec.) * N / 2 where A_TO is the timeout angle in electrical degrees, T_TO is the time period limit in seconds, ω is the rotor speed in rpm, and n is the number of rotor poles. Consequently, for a fixed timeout period, the corresponding timeout angle increases linearly with the speed of the rotor. Controller 16 can therefore use a fixed timeout period, T_TO. However, better control is possible if controller 16 uses a different timeout period for different rotor speeds. Controller 16, therefore, comprises a timeout lookup table that stores a timeout period, T_TO, for each of a plurality of rotor speeds.
In response to an edge of the Z_CROSS signal, controller 16 selects, from the timeout query table, a timeout turning period, T_TO, corresponding to the speed of rotor 5. The same timeout period is then used by the controller 16 until such time that an additional edge of the Z_CROSS signal is detected. In response to an additional edge of the Z_CROSS signal, controller 16 selects, from the timeout query table, a new timeout period corresponding to the speed of rotor 5. Therefore, as well as the advance period and the free rotation, the timeout period is updated only when the AC 4 supply voltage goes through zero, and remains constant during each AC 4 semiconductor cycle.
CONSTANT POWER
Controller 16 operates primarily in single switching, advanced switching mode. Within this mode, the speed of the rotor 5 varies as the rotor 5 experiences different loads. As the speed of the rotor varies, so does the
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19/57 magnitude of the electromotive force induced in the phase 7 winding. If controller 16 employed a fixed lead time and superheat period, the average input power and average output power of system 1 would vary with the speed of the rotor. However, there may be applications for which it is desirable to have a motor system 1 that maintains a constant average input or output power over a particular speed range.
The average input and output power of motor system 1 is also dependent on the RMS voltage of the AC supply 4. However, the RMS voltage may not be regular. Again, there may be applications for which it is desirable to have a motor system 1 that maintains a constant average input or output power, regardless of changes in AC supply voltage 4. Additionally, the mains supply in two different countries may vary as for the RMS voltage, but not the frequency. Therefore, it would be advantageous if the same performance were achieved by engine system 1 in both countries.
Consequently, in order to keep the average power constant (input or output) over a particular speed range and / or voltage range, controller 16 adjusts the advance period and the overheat period in response to changes in rotor speed and / or RMS voltage of the AC supply 4.
The advance lookup table, therefore, stores an advance period, T_ADV, for each of a plurality of rotor speeds and a plurality of stresses. Controller 16 also comprises an overheat offset query table that stores an overheat offset value, T_OVR_OFFSET, for each of a plurality of rotor speeds and a plurality of voltages. Each lookup table is, therefore, two-dimensional and is indexed by rotor speed and voltage. As described in more detail below, controller 16 samples the DC_SMOOTH signal to obtain a measurement of the RMS voltage of the AC supply 4, which is then used by controller 16 to index each of the lookup tables.
As in the lead-up period, controller 16 updates the overheat offset value in response to the edges of the Z_CROSS signal. In particular, controller 16 selects, from the overheat deviation query table, an overheat deviation value, T_OVR_OFFSET, corresponding to the rotor speed and the RMS voltage of the AC supply 4. The overheating deviation is therefore updated only when the AC 4 supply voltage goes through zero, and remains constant during each AC 4 semiconductor cycle.
In response to each edge of the HALL signal, controller 16 selects, from the superheat sine query table, a superheat sine value, T OVR SINE, corresponding to the time period, t, which has elapsed since the crossing
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20/57 at previous zero on AC power 4. Controller 16 then adds the superheat offset value, T_OVR_OFFESET, and the superheat sine value, T_OVR_SINE, to obtain the superheat period, T_OVR.
The lead time, T_ADV, and the superheat period, T_OVR, can therefore be defined as:
T_ADV = T_ADV_TABLE [speed, tension]
T_OVR = T_OFFSET_TABLE [speed, tension] + T_OVR_SINE_TABLE [t]
The advance period and the overheat period are thus adjusted in response to changes in the rotor speed and the RMS voltage of the AC 4 supply to ensure that the constant average power (input or output) is achieved. One could also adjust the overheating amplitude, T_OVR_AMP, in response to changes in rotor speed and / or RMS voltage. For example, controller 16 could store an overheat amplitude lookup table that stores an overheat amplitude value, T_OVR_AMP, for each of a plurality of rotor speeds and / or electrical voltages. Controller 16 would then update the overheat amplitude value in response to each edge of the Z_CROSS signal. The superheat sine value, T_OVR_SINE, would then be obtained by multiplying the superheat amplitude value, T_OVR_AMP, by the value obtained from the sine query table. However, multiplying two numbers increases the number of instructions executed by controller 16. In addition, a controller 16 having a higher bit resolution would be required in order to manage multiplication. Therefore, so that a relatively simple and inexpensive microcontroller can be used for controller 16, the overheating amplitude is not adjusted. However, if it is necessary or necessary to do this, the amplitude of overheating could also be adjusted.
The constant average power is maintained over a speed range limited by SPEED_CP_MIN and SPEED_CP_MAX, and over a voltage range limited by V_CP_MIN and V_CP_MAX. Outside these ranges, controller 16 does not attempt to keep the average power constant. The reasons for this may vary, depending on the particulars of the motor system 1. For example, at voltages below V_CP_MIN, it may not be possible to conduct sufficient current for the winding in phase 7 during each electric semicycle of motor 2 in order to maintain the constant average power. Alternatively, the efficiency of motor system 1 can drop considerably at voltages below V_CP_MIN, or the act of keeping the average power constant below that voltage can result in excessive current harmonics.
Although constant average power is maintained over a specific speed range and a voltage range, motor system 1 is nevertheless capable of operating effectively at speeds and voltages outside these ranges. Engine system 1,
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21/57 therefore, it has an operational speed range defined by SPEED_MIN and SPEED_MAX, and an operational voltage range defined by V_MIN and V_MAX. The advance lookup table and the superheat deviation lookup table store values that cover the entire operating speed and voltage ranges of the engine system 1 However, the constant average power is achieved only at speeds between SPEED_CP_MIN and SPEED_CP_MAX, and the voltages between V_CP_MIN and V_CP_MAX.
VOLTAGE AND TEMPERATURE MEASUREMENT
The peripherals 19 of the controller 16 include an analog-to-digital converter (ADC) with a plurality of channels. A first ADC channel is coupled to the input pin of the DC_SMOOTH signal, and a second ADC channel is coupled to the input pin of the TEMP signal.
In order to measure the RMS voltage of the AC supply 4, processor 17 selects the first channel of the ADC and samples the DC_SMOOTH signal using the ADC. The time constant of circuit R6, C2 is long enough for the DC_SMOOTH signal to appear relatively constant during each AC power semiconductor 4. The DC_SMOOTH signal therefore provides a measure of the peak voltage of the AC power 4. Once the peak voltage is directly proportional to the RMS voltage, DC_SMOOTH also provides a measure of the RMS voltage. Although the DC_SMOOTH signal is relatively constant during each AC power semiconductor 4, the signal nevertheless has a small degree of high frequency ripple that originates from the switching of inverter 10. Therefore, in order to compensate for this ripple, the processor 17 samples the DC_SMOOTH signal several times during each cycle of AC power 4. Processor 17 then averages the samples to obtain a measure of the RMS voltage of AC power 4.
In order to measure the temperature, processor 17 selects the second channel of the ADC and samples the TEMP signal using the ADC. Again, processor 17 samples the TEMP signal several times and averages it to obtain a temperature measurement. By averaging numerous samples, controller 16 does not inadvertently react to false noise in the TEMP signal or to temporary temperature changes within the motor system 1.
When operating at relatively high rotor speeds, the time required by the ADC to select a channel and sample the relevant input signal can delay the execution of other routines. As explained in more detail below, any delay is prone to affect the performance of engine system 1. Therefore, during the sampling of DC_SMOOTH or TEMP, the sampling process is divided into three distinct stages, each of which is performed, by turn in response to an edge of the HALL signal.
In response to a first edge of the HALL signal, processor 17 selects the appropriate ADC channel. In response to a second edge, processor 17 initiates the
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22/57
ADC. In response to a third edge, processor 17 reads the output record from the ADC. The output read by processor 17, therefore, represents a single sample of the selected input signal, i.e., DC_SMOOTH or TEMP. The sample read by processor 17 is then stored on memory device 18. This three-step process is then repeated to obtain an additional sample of the input signal, which is then added to the value already stored on memory device 18. The stored value in memory device 18, therefore, corresponds to the sum of individual samples read by processor 17. The three-step process is repeated for a predetermined number of times. The processor 17 then divides the value stored in the memory device 18 by the predetermined number to obtain an average measure of the input signal.
By dividing the sampling process into three distinct steps, the time required to sample the input signal is spread over three motor semiconductors 2. As a result, the time spent by the controller 16 in sampling the input signal during each motor semiconductor 2 is significantly reduced, and thus the possibility of event conflict is reduced.
Controller 16 samples the DC_SMOOTH and TEMP signals simultaneously. As illustrated in Figure 7, controller 16 performs the three-step process on the DC_SMOOTH signal to obtain a single sample of the RMS voltage from the AC supply. Controller 16 then performs the three-step process on the TEMP signal to obtain a single sample of the temperature. This process of alternately sampling DC_SMOOTH and TEMP is then repeated for a predetermined number of times. The memory device 18 therefore stores a first value corresponding to the sum of the samples of the RMS voltage of the AC supply 4, and a second value corresponding to the sum of the temperature samples.
Instead of sampling the DC_SMOOTH and TEMP signals simultaneously, controller 16 can then sample the two input signals sequentially. In particular, controller 16 can sample the DC_SMOOTH signal a predetermined number of times before sampling the TEMP signal one by a predetermined number of times. By sampling the two input signals sequentially instead of simultaneously, the ADC channel is changed only once for each set of samples. As a result, the step of selecting a channel can be eliminated from all samples except the first one. A two-step process (ie, starting ADC and reading ADC) can then be used to sample all but the first of the samples. Therefore, a larger number of samples can be collected over a particular period of time. However, a disadvantage of sampling the two input signals sequentially is that, for each signal, there is a period during which the signal is not being measured.
Controller 16 has only one ADC, which is required to sample two signals
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23/57 input, namely DC_SMOOTH and TEMP. It is for this reason that the sampling process includes the step of selecting an ADC channel. If the temperature sensor 15 were omitted from the control system 3, the RMS voltage of the AC supply 4 would be sampled without the need for channel selection. Alternatively, if peripherals 19 of controller 16 included an additional ADC, each input signal could be sampled by a dedicated ADC, and thus the step of selecting a channel could be omitted again. However, in both cases, the sampling process continues to be divided into two stages, so that the time required to sample an input signal is distributed between two motor semiconductors 2.
In the particular embodiment illustrated in Figure 7, each step of the sampling process is performed in response to a successive edge of the HALL signal. This then offers the advantage that each sample is obtained relatively quickly, that is, after three edges of the HALL signal. However, it is not essential that each step be performed in response to the successive edges of the HALL signal. For example, each step of the sampling process can be performed in response to each second or third edge of the HALL signal. Although this then requires a longer period of time to obtain each sample, controller 16 can use the time it is not dealing with the ADC to perform other routines.
Instead of using the average of the various samples as a measure of the input signal, controller 16 could alternatively use the sum of the samples. Alternatively, controller 16 could use the peak value of the samples as a measure of the input signal. For example, after reading a sample from the ADC output record, processor 17 could compare the sample to a value stored on memory device 18. If the sample is greater than the value stored on memory device 18, processor 17 overwrite the value with the sample. The comparison and rewrite steps are then repeated for each of the predetermined number of samples. After all samples have been collected, the value stored in memory device 18 represents the peak value of the samples. When measuring the peak value, it is not essential that the input signal representing the AC supply voltage 4 be uniform, as long as the samples span at least one semiconductor of the AC supply 4. Consequently, the smoothing capacitor C2 can be omitted or capacitance could be considerably reduced in order to reduce the size and / or cost of the control system 3.
In addition to measuring AC supply voltage 4 and temperature while operating at speed, controller 16 also measures voltage and temperature during startup. This initial check is carried out in order to ensure that the RMS voltage of the AC supply 4 and the temperature within the motor system 1 remain within the operational limits. During this initial stage, the time spent by controller 16 in sampling the input signal is not crucial. Consequently, during initialization, the controller
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24/57 sample the voltage and temperature without dividing the process into three stages.
CONSULTATION TABLES
The memory device 18 of the controller 16 stores a series of look-up tables, each having a particular speed and / or voltage resolution. The resolution of each lookup table need not be the same as the other lookup tables and may vary across the lookup table. For example, the advance lookup table can store an advance period every 10 krpm at relatively low speeds, which gradually increase up to 1 krpm at relatively high speeds.
Advanced switching is used in both multi-switching mode and single switching mode. In the multi-switching mode, the advance period is adjusted in response to changes in both the rotor speed and the RMS voltage of the AC 4 supply in order to keep the average power constant. In multi-switching mode, it is not necessary to adjust the lead time in response to changes in voltage. Therefore, in order to minimize the amount of memory required to store the feed lookup table, memory device 18 stores two feed lookup tables: a one-dimensional lookup table that is indexed by the speed of the rotor while operating at speeds below of SPEED_SINGLE, and a two-dimensional query table that is indexed by the speed of the rotor and the voltage at speeds equal to or above SPEED_SINGLE.
Instead of storing absolute values, each lookup table can instead store difference values. Controller 16 then stores a reference value to which the difference values are applied. Consequently, when updating a particular parameter, controller 16 indexes the relevant lookup table to select a different value, and applies the difference value to the reference value to obtain the parameter. So, for example, controller 16 can employ feed periods of 47 ps, 50 ps and 52 ps for the rotor speeds of 85 krpm, 90 krpm and 95 krpm. Controller 16 could then store 50 ps as the reference value. The advance lookup table would then store -2 ps, 0 ps and 1 ps for each of the three speeds. Storing a difference value usually requires less memory than an absolute value. As a result, lookup tables can be stored more efficiently. A higher resolution for the lookup tables could then be achieved for a given amount of memory. Alternatively or in addition, a cheaper controller with a smaller memory capacity can be used. In a more general sense, therefore, each lookup table can be said to store a control value (for example, an absolute or difference value) that is used by controller 16 to determine the relevant parameter, for example, lead time , overheating bypass, etc.
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25/57 controller 16 only updates the parameters that are required for the relevant operating mode. For example, while operating in synchronous switching mode, controller 16 has no need to select or update a lead time. Similarly, while operating in single switching mode, controller 16 has no need to select or update a free spin period. As a consequence of updating only the parameters required for a particular operating mode, controller 16 does not immediately switch from multi-switching mode to single switching mode when the rotor reaches SPEED_SINGLE. If controller 16 immediately switched from multi-switching mode to single switching mode, controller 16 would not know what period of time has elapsed since the previous zero crossing at AC supply voltage 4. As a result, controller 16 would not know which overheating period to use. Therefore, when the rotor speed reaches SPEED_SINGLE, controller 16 waits until the next edge of the Z_CROSS signal before switching from multi-switching mode to single switching mode.
As engine 2 accelerates from stationary speed, the duration of the Hall period decreases. Consequently, if a parameter (for example, the free spin period) was updated on each umpteenth edge of the HALL signal, the interval between each update would gradually decrease. Each parameter would then be updated less frequently at low speeds and more frequently at high speeds. By updating each parameter in response to a zero crossing in the AC 4 supply voltage, each parameter is updated at regular intervals, regardless of speed.
If a parameter were updated on each umpteenth edge of the HALL signal while operating in single switching mode, the parameter would be updated at different points within the AC 4 power cycle. This, in turn, could potentially increase the harmonic content of the current wave drawn from AC power 4. In addition, when motor system 1 is operating at a constant average speed, the instantaneous speed of rotor 5, however, varies during each AC power semiconductor 4 thanks to the sinusoidal increase and decrease DC link voltage. If a parameter were updated on each umpteenth edge of the HALL signal, a different value for the parameter could be selected despite the fact that the average speed of motor system 1 has not been changed. Again, this can result in an increase in harmonics within the current waveform drawn from AC power 4. By updating each parameter in response to a zero crossing on AC power 4, the same reference point in the AC power cycle 4 is used. Consequently, a more stable current waveform is obtained. Furthermore, by updating the parameters only once for each AC power semiconductor 4, the instructions executed by controller 16 are kept relatively simple, and thus, a simpler and cheaper microcontroller can be used. Without
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26/57 doubts, if desired, the various parameters can be updated less frequently by updating and each umpteenth edge of the Z_CROSS signal.
ENGINE OPERATION
The operation of motor system 1, as it accelerates from steady state to operating speed, will be described below. As can be seen in Figure 8, controller 16 has six operating modes: Startup, Stationary, Low Speed Acceleration, High Speed Acceleration, Operation and Failure. Within the various operating modes, controller 16 controls motor 2 through the use of one or more of the following four parameters: free spin period, advance period, overheat period and timeout period.
Startup mode
At startup, controller 16 activates peripherals 19 and samples the DC_SMOOTH signal and TEMP signal to obtain a measurement of the RMS voltage of the AC supply 4 and the temperature within the motor system 1. If the RMS voltage is less than a low voltage threshold, V_MIN, or greater than a high voltage threshold, V_MAX, controller 16 writes a “Low Voltage” or “High Voltage” error to memory device 18 and enters Fault Mode. Similarly, if the temperature is less than a lower temperature threshold, TEMP_MIN, or greater than a higher temperature threshold, TEMP_MAX, controller 16 writes a “Low Temperature” or “High Temperature” error to the memory device 18 and enter Failure Mode.
If the RMS voltage and temperature are within the operational thresholds, controller 16 determines whether the speed of rotor 5 exceeds a stationary threshold, SPEED_STATIONARY. As noted above, the speed of rotor 5 is obtained from the interval between two successive edges of the HALL signal, that is, the Hall period. If controller 16 fails to detect two edges of the HALL signal within a period of time corresponding to SPEED_STATIONARY, controller 16 enters Stationary Mode. Otherwise, controller 16 enters Low Speed Acceleration Mode.
Stationary Mode (ω <SPEED STATIONARY)
Controller 16 drives motor 2 in the reverse direction for a predetermined reverse drive time, T_RD. For the purposes of this description, we will assume that motor 2 is driven forward in response to triggering the phase 7 winding from left to right when the HALL signal is logically low and from right to left when the HALL signal is logically loud . Motor 2, therefore, is driven in the reverse direction in response to the triggering of the phase 7 winding from right to left when the HALL signal is logically low and from left to right when the HALL signal is logically high, as detailed in Figure 9.
Momentarily starting motor 2 in the reverse direction would cause the rotor 5 gi
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27/57 rasse or in a forward direction or adopt an angular position in relation to stator 6. Whether rotor 5 rotates forward or aligns with stator 6 will depend on the initial position of rotor 5. Rotor 5, therefore, is either moving in a forward direction or is in a position ready for acceleration in a forward direction.
After momentarily driving motor 2 in the reverse direction, controller 16 switches the winding in phase 7 in order to drive motor 2 forward. The forward drive should cause the rotor 5 to rotate in a forward direction. If rotor 5 is turning as expected, an edge of the HALL signal should occur within a predetermined time, T_FD. If no edges are detected within the predetermined time, T_FD, the controller writes a “Failed to Start” error to memory device 18 and enters Failure Mode. Otherwise, controller 16 switches the winding in phase 7 in response to the edge of the HALL signal in order to continue driving motor 2 forward. A second edge of the HALL signal should then occur within a period of time corresponding to SPEED_STATIONARY. If the second edge is detected within the predetermined time, controller 16 enters Low Speed Acceleration Mode. Otherwise, the controller writes a “Failed to Start” error to memory device 18 and enters Failure Mode.
Low Speed Acceleration Mode (SPEED STATIONARY <ω <SPEED ADV)
When operating in Low Speed Acceleration Mode, controller 16 drives motor 2 in multi-switching, synchronous switching mode. Figure 10 illustrates the waveforms of the HALL signal, the control signals and the phase current during some Hall periods.
In response to each edge of the HALL signal, controller 16 immediately switches the winding in phase 7 (i.e., inverting DIR1 and DIR2 and setting FREEWHEEL #). Controller 16 then determines the speed of rotor 5 based on the Hall period, T_HALL. Controller 16 then interrogates a zero crossing flag, which is set in response to an edge of the Z_CROSS signal. If the zero crossing flag is set and the speed of motor 5 is greater than or equal to SPEED_ADV, controller 16 enters High Speed Acceleration Mode. If, on the other hand, the zero crossing flag is set, but the rotor speed is less than SPEED_ADV, controller 16 updates the free spin period, T_FW, and the timeout period, T_TO, and releases the zero crossing flag. The free spin and timeout periods are updated by indexing the free spin and timeout query tables using the rotor speed.
After interrogating the zero crossing flag, and, if necessary, updating the free spin and timeout periods, controller 16 performs one of the three steps used
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28/57 to sample the DC_SMOOTH and TEMP signals. If the predetermined number of samples has been collected, controller 16 averages the samples to obtain a measure of the RS voltage of the AC supply 4 or the temperature within the motor system 1. If the RMS voltage is less than V_MIN or greater than V_MAX, or if the temperature is lower than TEMP_MIN or higher than TEMP_MAX, controller 16 writes a corresponding error to memory device 18 and enters Fault Mode.
After switching, controller 16 continues to excite the winding in phase 7 until an overcurrent event occurs or the timeout period, T_TO, expires. In response to either of these two events, controller 16 freely rotates the winding in phase 7 (ie, releasing FREEWHEEL #) for the free spin period, T_FW. If, at the end of the free spin period, the current in the phase 7 winding exceeds the overcurrent threshold, the controller 16 again freely turns the phase 7 winding for the free spin period, T_FW. Otherwise, at the end of the free spin period, controller 16 resumes the winding excitation in phase 7 (ie, setting FREEWHEEL #).
Controller 16 therefore switches the winding in phase 7 in sync with the edges of the HALL signal, and updates the free spin period and timeout period in response to the edges of the Z_CROSS signal. Controller 16 continues to drive motor 2 in synchronous switching multi-switching mode until the speed of rotor 5 reaches SPEED_ADV. Upon reaching SPEED_ADV, controller 16 enters High Speed Acceleration Mode in response to the next edge of the Z_CROSS signal.
High Speed Acceleration Mode (SPEED ADV <ω <SPEED SINGLE)
When operating in High Speed Acceleration Mode, controller 16 starts motor 2 in multi-switching, advanced switching mode. Figure 11 illustrates the waveforms of the HALL signal, the control signals and the phase current during some Hall periods.
In response to each edge of the HALL signal, controller 16 determines the speed of rotor 5 based on the Hall period, T_HALL. Controller 16 then interrogates the zero crossing flag, which is set in response to an edge of the Z_CROSS signal. If the zero crossing flag is set and the speed of motor 5 is greater than or equal to SPEED_SINGLE, controller 16 enters Travel Mode. If, on the other hand, the zero crossing flag is set and the rotor speed is less than SPEED_SINGLE, controller 16 updates the lead time, T_ADV, the free turn period, T_FW, and the timeout period , T_TO, and release the crossover flag at zero. The lead, free spin and timeout periods are updated by indexing the corresponding query tables using the rotor speed.
After interrogating the zero crossing flag, and, if necessary, updating the lead, free spin and timeout periods, controller 16 calculates the switching periodPetition 870190080086, of 8/19/2019, pg. 35/93
29/57 tion, T_COM, subtracting the advance period, T_ADV, from the Hall period, T_HALL. Controller 16 then loads the switching period, T_COM, into a timer.
After calculating the switching period, controller 16 performs one of the three steps used to sample the DC_SMOOTH and TEMP signals. If the predetermined number of samples has been collected, controller 16 averages the samples to obtain a measure of the RS voltage of the AC supply 4 or the temperature within the motor system 1. If the RMS voltage is less than V_MIN or greater than V_MAX, or if the temperature is lower than TEMP_MIN or higher than TEMP_MAX, controller 16 writes a corresponding error to memory device 18 and enters Fault Mode.
Controller 16 subsequently switches the winding in phase 7 (i.e., inverting DIR1 and DIR2 and setting FREEWHEEL #) after the timer has counted the switching period, T_COM. As a result, controller 16 switches the winding in phase 7 before the next edge of the HALL signal for the lead time, T_ADV. After switching, controller 16 excites the winding in phase 7 until an overcurrent event occurs or the timeout period, T_TO, expires. In response to either of these two events, controller 16 freely rotates the winding in phase 7 (ie, releasing FREEWHEEL #) for the free spin period, T_FW. If, at the end of the free spin period, the current in the phase 7 winding exceeds the overcurrent threshold, the controller 16 again freely turns the phase 7 winding for the free spin period, T_FW. Otherwise, at the end of the free spin period, controller 16 resumes the winding excitation in phase 7 (ie, setting FREEWHEEL #).
Controller 16, therefore, switches the winding in phase 7 before the edges of the HALL signal, and updates the lead time, free spin period and timeout period in response to the edges of the Z_CROSS signal. Controller 16 continues to drive motor 2 in the advanced switching multi-switching mode until the speed of rotor 5 reaches SPEED_SINGLE. Upon reaching SPEED_SINGLE, controller 16 enters Shift Mode in response to the next edge of the Z_CROSS signal.
Travel Mode (SPEED SINGLE <ω)
When operating in Travel Mode, controller 16 drives motor 2 in single switching, advanced switching mode. Figure 12 illustrates the waveforms of the HALL signal, the control signals and the phase current during some Hall periods.
In response to each edge of the HALL signal, controller 16 determines the speed of rotor 5 based on the Hall period, T_HALL. The speed of rotor 5 is expected to remain within a speed range limited by SPEED_MIN and SPEED_MAX. Controller 16 will, however, allow temporary speeds outside this range. Therefore, if the speed of rotor 5 drops below SPEED_MIN for a period of time longer than T US, controller 16 writes an error “Below Speed on memory device 18
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30/57 and enter Failure Mode. Similarly, if the speed of rotor 5 exceeds SPEED_MAX for a period of time longer than T_OS, controller 16 writes an error “Over Speed on memory device 18 and enters Fault Mode. However, if the speed of rotor 5 exceeds SPEED_TRIP, controller 16 immediately records a “Speed Trip” error on memory device 18 and enters Fault Mode. At speeds greater than SPEED_TRIP, the probability of mechanical and / or electrical failure increases considerably.
Controller 16 then interrogates the zero crossing flag, which is set in response to an edge of the Z_CROSS signal. If the zero crossing flag is set, controller 16 updates the lead time, T_ADV, and the overheat offset value, T_OVR_OFFSET. Each value is updated by indexing the relevant lookup table using the rotor speed and the measured RMS voltage of the AC supply 4. After updating the advance period and the overheat deviation value, controller 16 releases the zero crossing flag and starts a zero crossing timer.
After interrogating the zero crossing flag, and, if necessary, updating the lead time and the overheat deviation value, controller 16 calculates the switching period, T_COM, subtracting the lead period, T_ADV, from the Hall period, T_HALL. Controller 16 then loads the switching period, T_COM, into a timer. After determining the switching period, controller 16 indexes the superheat sine lookup table using the time stored by the zero crossing timer, t, to select a superheat sine value, T_OVR_SINE. Controller 16 then adds the superheat offset value, T_OVR_OFFESET, and the superheat sine value, T_OVR_SINE, to obtain the superheat period, T_OVR.
After determining the switching and overheating periods, controller 16 performs one of the three steps used to sample the DC_SMOOTH and TEMP signals. If the predetermined number of samples has been collected, controller 16 averages the samples to obtain a measure of the RS voltage of the AC supply 4 or the temperature within the motor system 1. If the RMS voltage is less than V_MIN or greater than V_MAX, or if the temperature is lower than T_MIN or higher than T_MAX, controller 16 writes a corresponding error to memory device 18 and enters Fault Mode.
Controller 16 subsequently switches the winding in phase 7 (i.e., inverting DIR1 and DIR2 and setting FREEWHEEL #) after the timer has counted the switching period, T_COM. As a result, controller 16 switches the winding in phase 7 before the next edge of the HALL signal for the lead time, T_ADV. After switching,
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31/57 controller 16 excites the winding in phase 7 until an overcurrent event occurs. In response to the overcurrent event, controller 16 continues to excite the winding in phase 7 for a period of overheating, T_OVR. After the overheating period has elapsed, controller 16 freely rotates the winding in phase 7 (ie, releasing FREEWHEEL #). The free spin then continues indefinitely until controller 16 then switches the winding in phase 7. Controller 16 therefore switches the winding in phase 7 before each edge of the HALL signal, updates the overheating period in response each edge of the HALL signal and updates the lead time and the overheat deviation value in response to each edge of the Z_CROSS signal.
When operating in Travel Mode, the magnitude of the electromotive force induced in the phase 7 winding by rotor 5 is of sufficient magnitude that the current in the phase 7 winding is unlikely to rise to an excessive level, even in the absence of a overcurrent. As a result, no timeout period is employed by controller 16 during operation in Travel Mode. This then reduces the number of instructions executed by controller 16.
Controller 16 drives motor 2 over a range of operating speed limited by SPEED_MIN and SPEED_MAX, the speed varying in response to changes in load. Within this speed range, controller 16 selects control values that ensure that the average constant power is achieved between SPEED_CP_MIN and SPEED_CP_MAX. Consequently, the constant average power is achieved for different loads. Controller 16 also drives motor 2 over a voltage range limited by V_MIN and V_MAX. Within this voltage range, controller 16 selects control values that ensure that the average constant power is achieved between V_CP_MIN and V_CP_MAX. Consequently, the same power and performance is achieved, regardless of variations in the voltage of the AC supply 4.
Fail mode
Controller 16 enters Fault Mode in response to an error, with the intention of preventing or limiting damage to motor system 1. Controller 16 therefore disables motor 2 by releasing DIR1 and DIR2 upon entering Fault Mode. Controller 16 may require that power to motor system 1 be turned off before motor system 1 can be reused. Alternatively, controller 16 can prevent any further use of motor system 1; this may depend on the type of failure that has occurred.
BENEFITS
For a conventional permanent magnet motor that is driven by an AC supply, the electromotive force induced in the phase winding makes it difficult to precisely control the amount of current drawn from the AC supply. As a result, forPetition 870190080086, of 8/19/2019, p. 38/93
32/57 ma of the current drawn from AC power will generally have a high harmonic content, resulting in an unsatisfactory power factor. In order to address this problem, conventional permanent magnet motors usually include an active power factor correction (PFC) circuit. The active PFC circuit emits a regular DC voltage for use in exciting the windings in phase, while ensuring that the current drawn from the AC supply is substantially sinusoidal. As a result, a relatively high power factor can be achieved. However, the inclusion of an active PFC circuit increases the cost of the engine system. In addition, the PFC circuit requires a high capacitance DC link capacitor so that the DC link voltage sampled by the PFC circuit is stable. Without a stable DC link voltage, the PFC circuit would estimate incorrect current demand levels, resulting in unsatisfactory current harmonics. However, a high capacitance DC link capacitor, in addition to being physically large, is also expensive.
With the motor system 1 of the present invention, the controller 16 employs an overcurrent threshold that is directly proportional to the DC link voltage and an overheating period that varies over each AC power semiconductor 4. The net result is that the controller 16 excites the winding in phase 7, during each electric semicycle of motor 2, for a driving period that varies throughout each semicycle of AC power 4. In particular, the duration of the driving period varies substantially as a semisinoid during each semicycle of AC 4 power. As a result, the waveform of the current drawn from AC 4 power approximates that of a sine wave. A relatively high power factor and a low harmonic content are therefore achieved without the need for a PFC circuit or a high capacitance bonding capacitor.
Figure 13 illustrates the current waveform that can be obtained with the motor system 1 of the present invention. The current waveform is superimposed with a perfect sine wave for comparison purposes. Using a driving period that varies over each AC 4 power semiconductor, a current waveform can be realized for which the amplitude of the lower order harmonics is relatively small. The high frequency ripple that can be seen in the current waveform in Figure 13 arises from the switching of the inverter.
A unit power factor is achieved for a current waveform with no harmonic content. As the harmonic content increases, the power factor decreases. The engine system 1 of the present invention, therefore, is capable of achieving a relatively high power factor. In fact, with the engine system 1 of the present invention, a power factor of at least 0.95 can be achieved. As a result, motor system 1 is able to achieve a relatively high average input power for a given peak current. On the other hand, a motor system with harmonics of a lower order related to Petition 870190080086, of 19/08/2019, p. 39/93
Large 33/57 will suffer from an unsatisfactory power factor. As a result, a lower average input power is achieved for the same peak current. In order to solve this problem, the level of the peak current could be increased. However, as the peak current increases, the efficiency of the system decreases due to increased energy losses. In addition, excessive peak currents can damage the inverter keys and / or demagnetize the rotor.
Many countries have laws that impose strict limits on the magnitude of current harmonics that can be extracted from the mains supply, for example, IEC61000-3-2. By using values suitable for the advance period, the overheating deviation and the overheating amplitude, the motor system 1 is able to meet the harmonic standards in all the operating speed and voltage ranges of the motor system 1. In fact, the current waveform in Figure 13, although not perfectly sinusoidal, complies with the harmonic standards established in IEC61000-3-2.
Unlike conventional motor systems, motor system 1 of the present invention achieves a current waveform with relatively small lower order harmonics without the need for an active PFC circuit or a high capacitance bonding capacitor. The connection capacitor C1 of the control system 3 is used only to smooth the relatively high frequency ripple that results from switching the inverter. The connection capacitor C1 does not need to smooth the DC connection voltage at the fundamental frequency. As such, a bonding capacitor can be used which results in a ripple at the DC bonding voltage of 50% or more at the fundamental frequency, ie, Vr = (Vmax-Vmin) / Vmax> 0.5. Controller 16, however, ensures that even at this level of ripple, the lower order harmonics are kept relatively small, and thus a satisfactory power factor can be achieved at a relatively high average input power. In fact, the current waveform in Figure 13 is achieved with a 100% voltage ripple. Since the C1 bonding capacitor only needs to filter out the high frequency switching ripple, a low-capacitance bonding capacitor can be used, thereby considerably reducing the cost and size of the motor system 1.
Thanks to the relatively high power factor that can be achieved with motor system 1, a relatively high average input power can be achieved despite the ripple of the DC link voltage. The average input power that can be achieved will, of course, depend on the RMS voltage of the AC supply 4. However, for an RMS voltage of 100 V, a constant average input power greater than 1000 W can be achieved, regardless of the ripple. DC link voltage. Consequently, when used with a mains supply, motor system 1 is capable of
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34/57 to achieve a constant average input power of at least 1000 W.
By selecting the appropriate values for the advance period, T_ADV, for the overheating deviation, T_OVR_OFFSET and for the overheating amplitude, T_OVR_AMP, a desired average input or output power can be achieved for motor system 1. In addition to the moreover, the appropriate values can be selected so that the efficiency of motor system 1 at each operating point (ie speed and voltage) is optimized for the desired input or output power. This means that several sets of values for T_ADV, T_OVR_OFFSET and T_OVR_AMP can result in the same desired average input or output power. However, from these various sets of values, one can select a single set that provides the best efficiency.
One or more of the advance period, the overheating deviation and the overheating amplitude can be adjusted in response to changes in rotor speed and / or in the RMs voltage of the AC 4 supply, so that a particular profile for the average power input or output is obtained over a speed range and / or voltage range. In particular, by adjusting at least the advance period and the overheating offset in response to changes in rotor speed and / or RMS voltage, the same average input or output power can be achieved.
For a single engine system, the appropriate values can be selected so that the variation in average power (input or output) does not exceed ± 1% over a speed range covering at least 10 krpm and / or a range of voltage spanning at least 10 V. However, if the same values are used in a plurality of mass-produced engine systems, the variation in average power for each engine system increases thanks to component and manufacturing tolerances. However, the appropriate values can be selected so that the variance in average power does not exceed ± 5% for a mass produced engine system over the aforementioned voltage and speed ranges. Constant average power (ie, within ± 5%) can also be achieved at relatively high speeds. In particular, the constant average power can be achieved over a speed range with a minimum value greater than 60 kpm and a maximum value greater than 80 krpm. In fact, constant average power can be achieved at speeds that exceed 100 krpm. In addition to achieving constant average power over a speed and / or voltage range, appropriate values can be selected so that an efficiency of at least 80% is maintained over the speed and / or voltage range.
The present invention, therefore, offers a high power motor system 1 that is capable of meeting existing harmonic standards without the need for an active PFC circuit or high capacitance bonding capacitor. Furthermore, engine system 1 is capable of achieving relatively high efficiency (ie at least 80%), as well
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35/57 as a constant average power (ie, within ± 5%) over a range of rotor speeds and RMS voltages.
CONFLICT OF EVENTS
Controller 16 performs different software routines in response to different events. For example, controller 16 performs a particular routine in response to an edge of the HALL signal. Controller 16 performs a different routine in response to an overcurrent event, and so on.
A relatively simple microcontroller typically includes a simple segmentation processor. Consequently, when the processor executes a routine in response to a particular event, the processor is unable to respond to other events until such time as it has finished executing the routine. Therefore, when the two events meet, the execution of one of the event routines will be delayed.
When operating at relatively low speeds6 of the rotor, any delay in executing a particular routine will be relatively short compared to the entire Hall period. Consequently, the delay does not tend to adversely affect the performance of engine system 1. Furthermore, the time spent at speeds below SPEED_SINGLE is expected to be relatively short, and therefore any effect that the event conflict may have on the performance of engine system 1 is not considered critical at these speeds. However, at speeds in SPEED_SINGLE or above, any delay in the execution of a routine can adversely affect the performance of motor system 1. In particular, the delay can affect one or more of the input energy, the output power, current efficiency and harmonics.
For example, when operating in single switching mode, controller 16 calculates the switching period, T_COM and the overheating period, T_OVR, in response to each edge of the HALL signal. If, during that time, an overcurrent event occurs, the overcurrent routine will not be executed until controller 16 has finished executing the Hall routine. As a result, more current than desired would be conducted to the phase 7 winding. Alternatively, if an edge of the HALL signal occurred while controller 16 is executing the overcurrent routine, the execution of the Hall routine would be delayed. Since the Hall routine is used to calculate the moment at which the phase 7 winding is switched, any delay in the execution of the Hall routine will have the effect of reducing the lead time. In each of these examples, the conflict tends to occur close to the zero crossing in the AC 4 supply voltage, thanks to the small overcurrent threshold. Consequently, despite the fact that the amount of current conducted to the phase 7 winding is not well controlled, the net effect on power and efficiency does not tend to be considerable. However, the net effect on current harmonics tends to be significant.
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Various measures can be taken to minimize the risk of event conflicts. In particular, the risk of conflict can be reduced by simplifying the instructions for each routine, so that the time required to perform each routine is kept relatively short. It is for this reason that controller 16 uses lookup tables that store control values in the form of time periods. By using query tables that store time periods, the mathematical calculations performed by the controller 16 can be kept relatively simple. In particular, mathematical calculations can be limited to simple addition (for example, when calculating the superheat period) and substrate (for example, when calculating the switching period). However, despite these measures, the event conflict can occur with a relatively simple processor at relatively high speeds.
The event conflict can be resolved by using a faster processor or multiple cores. However, both options increase the cost of controller 16. Therefore, two alternative schemes for driving motor 2 in single switching mode will be described below. Both schemes reduce the number of events that occur during each electric semicycle of engine 2 and thus reduce the possibility of event conflicts. Before describing the two alternative schemes, we will first take into account the events that occur during each motor 2 electric semicycle for the scheme described above. For the sake of clarity, the control scheme described above for single switching mode should hereinafter be called “single overcurrent switching mode”. The two alternative control schemes for the single switching mode will be called "single unlimited free-rotating switching mode" and "limited free-rotating single switching mode".
Single Overcurrent Switching Mode
A common method of event management is through the use of interrupts. In response to an interrupt, controller 16 interrupts the execution of the main code and performs the interruption by executing an interrupt service routine (ISR).
While operating in single overcurrent switching mode, controller 16 employs the following four interrupts. Hall, overcurrent, free rotation and switching. Figure 14 illustrates the waveforms of the HALL signal, the control signals and the phase current, as well as the interruptions employed by the controller 16 while operating in the single overcurrent switching mode.
The Hall interrupt is generated in response to an edge of the HALL signal. When serving the Hall interrupt, controller 16 first polls the zero crossing flag, which is defined in response to an edge of the Z_CROSS signal. If the zero crossing flag is set, controller 16 updates the advance period value and the overheat deviation value, and releases the zero crossing flag. The sign
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Z_CROSS, therefore, is used to set a flag instead of generating an interrupt. This then minimizes the total number of interruptions and, therefore, the possibility of interruption conflicts. After interrogating the zero crossing flag, controller 16 calculates the switching period, T_COM, and the overheating period, T_OVR. Controller 16 then loads the switching period into a first timer, Timer1. Finally, controller 16 performs one of the three steps used to sample the DC_SMOOTH and TEMP signals.
The overcurrent interruption is generated in response to a logically low overcurrent signal emitted by the current regulator 22. When serving the overcurrent interruption, controller 16 loads the overheating period, T_OVR, into a second timer, Timer2.
The free rotation interruption is generated by the second timer when the overheating period has elapsed. When serving the free rotation interruption, the controller 16 freely rotates the winding in phase 7.
The switching interruption is generated by the first timer when the switching period has elapsed. When serving the switching interruption, the controller 16 switches the winding in phase 7.
Since the overcurrent ISR is responsible for charging the second timer with the overheating period, it is not possible for the overcurrent and free spin interruptions to conflict. In addition, by ensuring that the lead time is longer than the time required to perform the switching ISR, it is possible to avoid the conflict of Hall and switching interrupts. However, four possible interruption conflicts are still possible, namely, Hall and overcurrent, Hall and free spin, switching and overcurrent, and switching and free spin.
Unlimited Free Spin Single Switching Mode
When operating in single unlimited free-spin switching mode, the overcurrent interruption is deactivated, that is, controller 16 ignores the overcurrent signal emitted by current regulator 22. In response to an edge of the HALL signal, controller 16 calculates a driving period, T_CD, in addition to the switching period, T_COM. Controller 16 switches the winding in phase 7 at a time, T_COM, after the edge of the HALL signal. After switching, controller 16 excites the winding in phase 7 during the driving period, T_CD, after which controller 16 freely rotates the winding in phase 7.
The driving time is similar to the overheating period used in the single overcurrent switching mode. In particular, the driving period comprises a deviation value and a sine value. However, unlike the overheating period, the conduction period waveform includes a phase shift in
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38/57 with respect to the AC supply voltage cycle 4.
In single overcurrent switching mode, controller 16 initially excites the winding in phase 7 until the current in the winding in phase 7 reaches an overcurrent threshold. Then, controller 16 excites the winding in phase 7 for a period of overheating, T_OVR. The total conduction period during which the phase 7 winding is excited is therefore the sum of the initial excitation period and the overheating period. The current in the winding in phase 7 is detected by monitoring the voltage through each of the bypass resistors R1, R2. More specifically, the voltage across each of the shunt resistors R1, R2 is output to controller 16 as a current detection signal, I_SENSE_1 and I_SENSE_2. As Figure 2 illustrates, each current detection signal is filtered by an RC filter R8, C3 and R9, C4, which acts to remove high-frequency noise. The RC filter time constant introduces a time delay between the measured current and the actual current in the phase 7 winding. The net result is that the conduction period waveform undergoes a phase shift in relation to the supply cycle AC 4. This phase shift helps to reduce the magnitude of the lower order current harmonics.
In the single unlimited free-spin switching mode, the overcurrent interruption is deactivated. The RC filters R8, C3 and R9, C4, therefore, do not influence the waveform of the driving period. Therefore, in order to replicate the phase shift that exists in the single overcurrent switching mode, the driving period waveform includes a phase shift in relation to the AC power cycle 4. The driving period, T_CD, therefore , is defined by the equation:
T_CD = T_CD_OFFSET + T_CD_AMP * abs {sen (Θ + A_CD_PHASE)} where T_CD_ OFFSET is a deviation value, T_CD_AMP * abs {sen (Θ + A_CD_PHASE)} is a rectified sine wave with an amplitude defined by T_CD_AMP. Θ is the angle in the voltage cycle of the AC supply 4 and A_CD_PHASE is a phase angle.
Both the angle Θ and the conducting phase angle, A_CD_PHASE, can be expressed as time intervals:
Θ (degrees) = t (sec) * f (Hz) * 360 (degrees)
A_CD_PHASE (degrees) = T_CD_PHASE (sec) * f (Hz) * 360 (degrees)
Consequently, the driving period can be defined as:
T_CD = T_CD_OFFSET +
T_CD_AMP * abs {sen ({t + T_CD_PHASE} * f * 360 degrees)}
For simplicity, the driving period, T_CD, can be considered as:
T_CD = T_CD_OFFSET + T_CD_SINE where T_CD_OFFSET is a conduction deviation value that is independent of
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39/57 time, and T_CD_SINE is a conduction sine value that is time dependent.
T_CD_SINE is stored by the controller 16 as a conduction sine query table, which comprises a conduction sine value, T_CD_SINE, for each of a plurality of times.
In single overcurrent switching mode, controller 16 adjusts the advance period and the overheat offset value in response to changes in rotor speed and AC supply voltage 4 to keep the average power constant. Similarly, in the single unlimited free-spin switching mode, controller 16 adjusts the lead time, T_ADV, and the lead offset value, T_CD_OFFSET, in response to changes in rotor speed and voltage to maintain the constant average power. Controller 16, therefore, stores an advance lookup table and a conduction lookup table, each of which is indexed by the rotor speed and voltage:
T_ADV = T_ADV_TABLE [speed, tension]
T_CD_OFFSET = T_CD_OFFSET_TABLE [speed, tension]
In single overcurrent switching mode, controller 16 initially excites the winding in phase 7 until the current in the winding in phase 7 reaches an overcurrent threshold. The overcurrent threshold is proportional to the DC link voltage, and thus the duration of this initial excitation period is sensitive to changes in the AC supply voltage 4. The duration of the initial excitation period is also sensitive to changes in the magnitude of the force. against electromotive induced in the phase 7 winding by rotor 5. Consequently, the initial excitation period is sensitive to changes in both the rotor speed and the AC supply voltage 4. Thanks to the RC filter that operates on each of the current, this initial excitation period introduces a phase delay in the conduction period waveform, which helps to reduce the magnitude of the lower order current harmonics. In a single unlimited free-spin switching mode, the phase delay is replicated by making the phase shift of the driving period waveform in relation to the voltage waveform of the AC power 4. Once the phase delay is sensitive to changes in rotor speed and AC supply voltage 4, controller 16 adjusts the phase of the conduction waveform in response to changes in rotor speed and voltage. The controller 16, therefore, comprises a conduction phase shift query table that stores a phase shift value, T_CD_PHASE_SHIFT, for each of a plurality of rotor speeds and a plurality of voltages. The driving period can therefore be defined as:
T_CD = T_CD_OFFSET_TABLE [speed, tension] +
T CD SINE TABLE [t + T CD PHASE SHIFT [speed, tension]]
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The lead time, T_ADV, the lead offset value, T_CD_OFFSET, and the lead phase offset value, T_CD_PHASE_SHIFT, are updated in response to the edges of the Z_CROSS signal. The values are therefore updated only when the voltage of the AC 4 supply goes through zero, and remain constant during each semicycle of the AC 4 supply.
In response to an edge of the HALL signal, controller 16 determines the time period, t, that has elapsed since the last edge of the Z_CROSS signal. Controller 16 then indexes the conduction sine lookup table using the sum of the elapsed time, t, and the conduction phase offset value, T_CD_PHASE_SHIFT, in order to select a conduction sine value, T_CD_SINE. Controller 16 then adds the conduction offset value, T_CD_OFFSET, and the conduction sine value, T_CD_SINE, to obtain the conduction period, T_CD.
Controller 16 employs these interrupts while operating in single, unlimited free-spin switching mode. Hall, free rotation and switching. Figure 15 illustrates the waveforms of the HALL signal, the control signals and the phase current, as well as the interruptions employed by the controller 16 while operating in the single unlimited free-rotating switching mode.
The Hall interrupt is generated in response to an edge of the HALL signal. When serving Hall interrupt, controller 16 interrogates the zero crossing flag, which is defined in response to an edge of the Z_CROSS signal. If the zero crossover flag is set, controller 16 updates the lead time, the lead offset value and the lead phase offset value, and releases the zero crossover flag. After interrogating the zero crossing flag, controller 16 calculates the switching period, T_COM, and the driving period, T_CD. Controller 16 then loads the switching period into a first timer, Timer1. Finally, controller 16 performs one of the three steps used to sample the DC_SMOOTH and TEMP signals.
The switching interruption is generated by the first timer when the switching period has elapsed. When serving the switching interrupt, controller 16 switches the winding in phase 7 and loads the driving period into a second timer, Timer2.
The free rotation interruption is generated by the second timer when the driving period has elapsed. When serving the free rotation interruption, the controller 16 freely rotates the winding in phase 7.
In comparison with the single overcurrent switching mode, controller 16 employs one less interruption. Furthermore, since the switching ISR is responsible for charging the second timer with the overheating period, it is not possible for switching and free-spin interruptions to conflict. Consequent
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41/57 the risk of interruption conflicts is reduced considerably.
By ensuring that the lead time is longer than the time required to serve the switching interruption (T_COM_ISR) and less than the Hall period minus the time required to serve the Hall interruption (T_HAL_ISR), it is possible to avoid the conflict of Hall interruptions and switching, ie
T_COM_ISR <T_ADV <T_HALL - T_HALL_ISR
However, it is still possible for the free spin interruption to conflict with the Hall interruption. However, as will be described below, controller 16 can be configured so that the free spin is initiated at the end of the driving period, without the need to generate an interruption.
There are known microcontrollers that have a timer capable of operating in the output comparison mode. In the output comparison mode, a comparator compares the timer counter record with an output comparison record. When the values of the two registers correspond, the comparator generates an interruption or defines / releases / activates a microcontroller output pin. The particular action taken by the comparator is typically defined by means of a register bit.
In one embodiment, the output comparison mode is exploited by controller 16 to release the FREEWHEEL # signal if an interrupt is generated. As Figure 16 illustrates, the peripherals 19 of the controller 16 comprise two timers 30,31 and a comparator module 32. The memory device 18 comprises a timer register 33, 34 for each of the timers 30, 31 and a comparison register 35. The first timer 30 is used to calculate the switching period, T_COM, and the second timer 31 is used to calculate the driving period, T_CD. The second timer 31 is configured to operate in the output comparison mode. Consequently, when serving the switching interruption generated by the first timer 30, the controller 16 switches the winding in phase 7, loads the driving period in the comparison record 35, and restarts the second timer 31. Comparator module 32 then compares the second timer record 34 and comparison record 35. When the two records 34, 35 correspond (which occurs when the driving period has already elapsed), comparator module 32 restarts an SR 36 hitch, which in turn releases a output pin 21 of controller 16. This output pin 21 is then used by controller 16 for the FREEWHEEL # signal. Therefore, when the driving period has elapsed, FREEWHEEL # is released without an interruption being generated. Once the output pin is engaged, FREEWHEEL #, continues to be released until controller 16 defines engagement 36 while serving the switching interruption.
Controller 16, therefore, is capable of operating in single, unlimited free-rotating switching mode using only two interrupts, namely, Hall and switching. However, with
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42/57 As noted above, the conflict of Hall and switching interruptions can be avoided by ensuring that the lead time is kept within particular limits. Consequently, the interrupt conflict can be avoided completely.
Depending on the type of microcontroller that is used for controller 16, it may not be possible to use the output comparison mode to calculate the driving period and release the output pin used for FREEWHEEL #. For example, the microcontroller may not have a timer capable of operating in the output comparison mode. Alternatively, the microcontroller can have an 8-bit timer and a 16-bit timer, and only the 16-bit timer is capable of operating in the output comparison mode. However, since the switching period is typically longer than the driving period, it may be necessary to use the 16-bit timer for the switching period. In cases for which the output comparison mode is not available to release FREEWHEEL #, a PWM module can be used in place to release FREEWHEEL # without resorting to an interrupt, as will now be described.
Figure 17 illustrates an alternative embodiment in which the peripherals 19 of the controller 16 comprise two timers 30,31 and a PWM module 37. The memory device 18 comprises a timer register 33, 34 for each of the timers 30, 31, a duty cycle record 38 and period record 38. The first timer 30 is again used to calculate the switching period, T_COM. The second timer 31, however, is used as a clock signal for the PWM 37 module. The PWM 37 module comprises a pair of comparators 40, 41 and an SR 42 coupling. A first comparator 40 compares the second timer register 34 and the duty cycle record 38. When the values of the two records 34, 38 match, the first comparator 40 restarts the SR 42 coupling, which in turn releases an output pin 21 from controller 16. A second comparator 41 compares the second timer record 34 and period record 39. When the values of these two records 34, 39 match, two things happen. First, the second comparator 41 defines the coupling SR 42, which, in turn, defines the output pin 21. Second, the second timer 31 is reset. Output pin 21 of controller 16 is therefore released when the second timer record 34 and duty cycle record 38 match, and is defined when the second timer record 34 and period record 39 match.
Output pin 21 activated by PWM module 37 is used by controller 16 for the FREEWHEEL # signal. During the service of the switching interruption, controller 16 switches the winding in phase 7, loads the driving period in the duty cycle record 38 and loads the second timer record 34 with the same value as the period record 39. In response, PWM module 37 defines FREEWHEEL # and the second
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43/57 timer record 34 is reset. The second timer 31 then increments the second timer record 34 until the second timer record 34 and duty cycle record 38 match. When the two records 34, 38 match (which occurs when the driving period has elapsed), the PWM 37 module releases FREEWHEEL #. The phase 7 winding is therefore freely rotated without the need for an interruption.
If the period record 39 is set too low, the second timer record 34 and the period record 39 can match during the free spin period. This would cause the FREEWHEEL # signal to be set prematurely. Therefore, so that the free spin is not completed before switching, the period record 39 stores a maximum possible value (for example, an 8 bit period record stores 0xFF).
In each of the above embodiments, a comparator compares the value of the second timer record with a value stored in a comparison record. When the second timer register and the comparison register match, the comparator releases the output pin used for FREEWHEEL #. In the first embodiment, the comparator forms part of a comparator module 32, while in the second embodiment, the comparator forms part of a PWM 37 module. However, any comparator in controller 16 can be used, as long as the comparator is capable of control (either directly or through the associated hardware) an output pin of the controller 16 in response to a comparison of the timer register and a comparison register.
Limited Free Spin Single Switching Mode
As with single unlimited free-spin switching mode, overcurrent interruption is deactivated during operation in single limited free-spin switching mode. In response to an edge of the HALL signal, controller 16 calculates a start-up period, T_DOFF, in addition to the switching period, T_COM. Controller 16 then continues to excite the phase 7 winding during the start-up period, T_DOFF, after which controller 16 freely spins the phase 7 winding. The free rotation then continues until controller 16 switches the phase 7 winding. .
The start-up period is similar to the driving period used in the single unlimited free-shift mode. In particular, the start-up period, T_DOFF, is defined by the equation:
T_DOFF = T_DOFF_OFFSET + T_DOFF_AMP * abs {sen (Θ + A_DOFF_PHASE)} where T_DOFF_ OFFSET is a deviation value and T_DOFF_AMP * abs {sen (Θ + A_DOFF_PHASE)} is a rectified sine wave with an amplitude defined by T_DOFF_AMP. Θ is the angle in the voltage cycle of the AC supply 4 and A_DOFF_PHASE is a phase angle.
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Both the angle θ and the start phase angle, A_DOFF_PHASE, can be expressed as time intervals:
θ (degrees) = t (sec) * f (Hz) * 360 (degrees)
A_DOFF_PHASE (degrees) = T_ DOFF _PHASE (sec) * f (Hz) * 360 (degrees)
Consequently, the start-up period can be defined as:
T_DOFF = T_DOFF_OFFSET +
T_DOFF_AMP * abs {sen ({t + T_OVR_PHASE} * f * 360 degrees)}
For simplicity, the start period, T_DOFF, can be thought of as: T_DOFF = T_DOFF_OFFSET + T_DOFF_SINE where T_DOFF_OFFSET is a start offset value that is independent of time, and T_ DOFF _SINE is a start sine value that is dependent of time.
The start-up period, T_DOFF, is stored and updated by controller 16 in the same way as described above for the driving period, T_CD. In particular, controller 16 stores a start sine lookup table that is indexed by time, and a start deviation lookup table and a start phase lookup table, each of which is indexed by speed of the rotor and the voltage of the AC supply 4. The start-up period can therefore be defined as:
T_DOFF = T_DOFF_OFFSET_TABLE [speed, tension] + T_DOFF_SINE_TABLE [t + T_DOFF_PHASE_SHIFT [speed, tension]]
Controller 16 employs these interrupts while operating in single limited free-spin switching mode. Hall, free rotation and switching. Figure 18 illustrates the waveforms of the HALL signal, the control signals and the phase current, as well as the interruptions employed by the controller 16 while operating in the single limited free-spin switching mode.
The Hall interrupt is generated in response to an edge of the HALL signal. When serving Hall interrupt, controller 16 interrogates the zero crossing flag, which is defined in response to an edge of the Z_CROSS signal. If the zero crossing flag is set, controller 16 updates the lead time, the start offset value and the start phase offset value, and releases the zero crossing flag. After interrogating the zero crossing flag, controller 16 calculates the switching period, T_COM, and the starting period, T_DOFF. Controller 16 then loads the switching period into a first timer, Timer1, and the starting period into a second timer, Timer2. Finally, controller 16 performs one of the three steps used to sample the DC_SMOOTH and TEMP signals.
The free spin interruption is generated by the second timer when the start-up period has elapsed. When serving the free rotation interruption, the controller 16 freely rotates the winding in phase 7.
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The switching interruption is generated by the first timer when the switching period has elapsed. When serving the switching interruption, the controller 16 switches the winding in phase 7.
Consequently, as in the single unlimited free-spin switching mode, controller 16 employs only three interrupts. This is different from the four interrupts used in single overcurrent switching mode. In addition, by ensuring that the lead time is longer than the time required to serve the switching interruption, it is possible to avoid the conflict of the Hall and switching interruptions.
As described above for the single unlimited free-spin switching mode, controller 16 can be configured so that the free spin is initiated at the end of the start-up period, without the need to generate an interruption. For example, the second timer can be configured to operate in the output comparison mode, so that the output pin used for FREEWHEEL # is released when the start-up period has elapsed. Alternatively, controller 16 can include a PWM module, which is used to activate the output pin for FREEWHEEL #. For example, during Hall interrupt service, controller 16 could load the communication period on the first timer, load the start-up period in the duty cycle record and restart the second timer. When the server subsequently interrupts the switching, controller 16 then switches the winding in phase 7 and loads the counter register of the second timer with the period register value.
Controller 16 can therefore be configured to operate in single limited free-spin switching mode using only two interrupts, namely, Hall and switching. However, as already noted, the conflict of these two interrupts can be avoided by ensuring that the lead time is longer than the time required to serve the switching interruption. Consequently, the interrupt conflict can be avoided completely.
In the limited free-spin single switching mode, the start-up period is referenced to an edge of the HALL signal. As a result, the free spin of the winding in phase 7 cannot begin until after the edge of the HALL signal. In the single unlimited free-spin switching mode, the driving period is referenced in relation to the switching. Since switching occurs before the edge of the HALL signal, the free rotation of the winding in phase 7 can begin before, during or after the edge of the HALL signal. It is for this reason that the two schemes are called limited free spin and unlimited free spin.
In comparison with the single overcurrent switching mode, both the unlimited and limited free-spin single switching modes employ fewer interruptions, and therefore the risk of interrupt conflict is reduced. In fact, controller 16 can be configured so that interrupt conflict is avoided entirely
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46/57 te. However, despite the potential for interrupt conflict, the single overcurrent switching mode has the advantage of self-compensating for tolerances and limitations within motor system 1. For example, controller 16 employs a zero crossover timer to monitor time elapsed since a zero crossing at the AC supply voltage 4. However, the zero crossing timer is started only as part of the Hall routine. There is, therefore, a variation in the time used to index the sine lookup table. In another example, there may be a tolerance in the balance of the activity cycle of the HALL signal. Any imbalance in the activity cycle will introduce an error in the Hall period. Therefore, tolerances and limitations in motor system 1 can result in small errors in the timing of certain events (eg switching, free spin, etc.). In single overcurrent switching mode, controller 16 initially excites the winding in phase 7 until the current in the winding in phase 7 reaches an overcurrent threshold. The duration of this initial excitation period is not controlled by controller 16. As a result, the initial excitation period acts to compensate for certain timing errors. Consequently, a more stable current waveform can be obtained when operating in single overcurrent switching mode. In addition to self-compensating for the tolerances and limitations within motor system 1, the initial excitation period introduces a phase delay that acts by attenuating the lower order current harmonics. In the limited and unlimited free-spin single-switching modes, this phase delay is replicated through the use of a phase-shift lookup table, which consumes valuable memory resources. A cheaper microcontroller with less memory, therefore, can be used for controller 16 while operating in single overcurrent switching mode. Alternatively, the memory that would be used for a phase shift lookup table could be used to improve the resolution of the other lookup tables, for example, the forward sine, overheat, or overheat lookup tables.
DRIVING PERIOD
In each of the three schemes described above, controller 16 excites the winding in phase 7 for a driving period, T_CD, in each electric semicycle of motor 2.
In the single overcurrent switching mode, the driving period, T_CD, can be defined as:
T_CD = T_OC + T_OVR where T_OC is the time it takes for the current in the winding in phase 7 to reach the overcurrent threshold, and T_OVR is the period of overheating. Consequently, the driving period, T_CD, can be defined as:
T CD = T OC + T OVR OFFSET + T OVR SINE
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The overcurrent threshold is directly proportional to the DC link voltage and thus varies as a rectified sinusoid. The current in the phase 7 winding rises at substantially the same rate, regardless of the level of the DC link voltage; the reasons for this behavior are outside the scope of this document. Consequently, the time it takes for the current in the phase 7 winding to reach the overcurrent threshold, T_OC, varies substantially as a semi-sinusoid during each AC power semiconductor 4. However, thanks to the RC filter acting constant on each of the current detection signals, the T_OC waveform is shifted in phase in relation to the voltage waveform of the AC supply 4.
The superheat deviation, T_OVR_OFFSET, is constant while the superheat sine value, T_OVR_SINE, varies as a semi-sinusoid during each AC power semiconductor 4. In addition, the waveform of the superheat sine value is in phase with the voltage waveform of AC power 4.
Since the overheating deviation is constant during each AC 4 semiconductor cycle, the variation in the driving period is defined by the sum of two semi-sine components, T_OC and T_OVR_SINE. The phase difference between the two components, which originates from the RC filter, is relatively small. In addition, the amplitude of T_OC is greater than that of T_OVR_SINE. Consequently, despite the phase difference, the sum of the two components resembles a rectified sinusoid having a phase shift in relation to the voltage waveform of the AC supply 4.
The length of the driving period, T_CD, therefore, varies as a periodic waveform. The waveform can be defined as the sum of two components: a first component (T_OVR_OFFSET) that is constant throughout each cycle of the waveform and a second component that varies (T_OC + T_OVR_SINE) during each cycle of the waveform . Each cycle of the waveform is repeated with each semicycle of AC 4 power. However, the waveform of the driving period undergoes a phase shift in relation to the voltage waveform of AC 4 power.
In the single unlimited free-spin switching mode, the driving period, T_CD, is defined by:
T_CD = T_CD_OFFSET + T_CD_SINE
The conduction deviation value, T_CD_OFFSET, is constant while the conduction sine value, T_CD_SINE, varies as a semisenoid during each AC power semiconductor 4. Furthermore, the conduction sine value waveform is deviated phase in relation to the voltage waveform of the AC supply 4. In fact, the phase shift is intended to replicate the phase shift that arises from the RC filter in single overcurrent switching mode.
Consequently, as in the single overcurrent switching mode, duration Petition 870190080086, of 8/19/2019, p. 54/93
The driving period varies as a periodic waveform. The waveform can again be defined as the sum of two components: a first component (T_CD_OFFSET) that is constant during each cycle of the waveform and a second component (T_CD_SINE) that varies during each cycle of the waveform. Each cycle of the waveform is repeated with each semicycle of the AC 4 supply, and the waveform has its phase shifted in relation to the voltage waveform of the AC 4 supply.
In the limited free-spin mode, the driving time, T_CD, can be defined as:
T_CD = T_ADV + T_DOFF where T_ADV is the lead period and T_DOFF is the start period. Consequently, the driving period, T_CD, can be defined as:
T_CD = T_ADV + T_DOFF_OFFSET + T_DOFF_SINE
The lead time, T_ADV, and the start deviation, T_DOFF_OFFSET, are constant while the start sine value, T_DOFF_SINE, varies as a semisenoid across each AC power semiconductor 4. Again, to reflect the deviation from phase originating from the RC filter in single overcurrent switching mode, the waveform of the start sine value undergoes a phase shift from the voltage waveform of the AC supply 4.
Consequently, as in the other two single switching modes, the length of the driving period varies as a periodic waveform. The waveform can be defined by the sum of two components: a first component (T_ADV + A_DOFF_OFFSET) that is constant during each cycle of the waveform and a second component (T_DOFF_SINE) that varies during each cycle of the waveform. Again, each cycle of the waveform is repeated with each semicycle of the AC 4 supply, and the waveform of the driving period has its phase shifted in relation to the voltage waveform of the AC 4 supply.
In each of the three schemes, the duration of the driving period is defined by a periodic waveform that is repeated with each AC 4 power cycle. More particularly, the waveform varies substantially as a semisenoid throughout each cycle of the cycle. wave shape. Consequently, the benefits described above in connection with the single overcurrent switching mode apply equally to the unlimited and limited single switching modes.
In each scheme, the driving period waveform is adjusted in response to changes in rotor speed 5 and / or the RMS voltage of the AC supply 4 in order to achieve a particular performance. For example, the deviation of the waveform is adjusted primarily so that the constant average power (or a particular profile for average power) is achieved over a range of speeds and / or voltages. THE
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49/57 phase of the waveform is adjusted primarily so that the magnitude of the lower order harmonics in the current waveform is kept below predetermined thresholds. The length of the driving period can be expressed as the sum of two components: a first component that is constant during each cycle of the waveform and a second component that varies during each cycle of the waveform. In response to changes in rotor speed and / or RMS voltage, the first component is adjusted to keep the average power constant and the second component is adjusted to maintain relatively small harmonics.
Although three different schemes for the single switching mode have been described, controller 16 is not necessarily limited to just one of these schemes when operating in single switching mode. Instead, controller 16 can use one or more of the three schemes when operating in single switch mode. For example, controller 16 may initially employ single overcurrent switching mode when the rotor speed reaches SPEED_SINGLE. As already noted, the single overcurrent switching mode has the benefit of offering a degree of self-compensation. However, as the rotor 5 accelerates, the Hall period shortens and, therefore, the risk of interruption conflict increases. Consequently, when the speed of the rotor reaches a predetermined threshold, controller 16 can switch from single overcurrent switching mode to single unlimited free-rotating switching mode.
For each of the three single switching schemes, the values for the advance period, the deviation, the amplitude and the phase deviation are obtained from the simulation. The simulation refines the various values for each operating point (for example, speed and voltage) in order to obtain the best performance (for example, better efficiency and / or lower order harmonics at the desired average input or output power.
SPECIFIC EXAMPLE
A particular embodiment of the engine system 1 will now be described by way of example only. The values for the various hardware components of motor system 1 are detailed in Figure 19, while Figure 20 lists several constants and thresholds employed by controller 16. Figures 21 and 22 detail the flow connection characteristics of the connection inductor L1 and engine 2.
As illustrated in Figure 23, engine system 1 has seven modes of operation: Fault, Startup, Stationary, Low Speed Acceleration, Medium Speed Acceleration, High Speed Acceleration and Travel. Therefore, in comparison to that described and illustrated earlier in Figure 8, engine system 1 has an additional mode of operation.
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50/57 Medium corresponds to the High Speed Acceleration Mode described above. Consequently, when operating in Medium Speed Acceleration Mode, controller 16 starts motor 2 in multi-switching, advanced switching mode.
When operating in High Speed Acceleration Mode, controller 16 starts motor 2 in single switching, advanced switching mode. The time spent by engine system 1 in High Speed Acceleration mode is relatively short. Therefore, in order to conserve memory, controller 16 does not store an overheat offset query table and an overheat sine query table. Instead, controller 16 stores a single superheat lookup table comprising an overheat period, T_OVR, for each of a plurality of rotor speeds. Controller 16 then updates the overheat period, along with the lead time, in response to the edges of the Z_CROSS signal. Consequently, the overheating period employed by controller 16 is constant during each AC power semiconductor 4. However, the use of a constant overheating period does not adversely affect the performance of motor system 1 for two reasons. First, the amount of time spent in High Speed Acceleration mode is relatively short. Second, the controller 16 initially excites the winding in phase 7 until the current in the winding in phase 7 exceeds a threshold that is proportional to the DC link voltage. Consequently, despite using a period of constant overheating during each semicycle of the AC 4 supply, the current waveform continues to approximate that of a sine wave. The lead time and the superheat period are updated in response to changes in rotor speed only and are not updated in response to changes in the RMS voltage of the AC supply. This then reduces the size of the lookup tables, thereby freeing up more memory for the most important tables used in Scroll Mode. Controller 16 continues to drive motor 2 in single overcurrent switching mode, advanced switching, until the moment when the rotor speed reaches SPEED_UFW. Upon reaching SPEED_UFW, controller 16 enters Operation Mode in response to the next edge of the Z_CROSS signal.
The Travel Mode corresponds to that described previously, but with one exception. Instead of employing the single overcurrent switching mode, controller 16 instead employs the unlimited free-spin single switching mode. Consequently, in addition to updating the lead time and offset value, controller 16 also updates the phase offset value in response to each edge of the Z_CROSS signal. Other than that, the operation of the controller 16 is essentially identical to that described above. In particular, controller 16 drives motor 2 over an operational speed range limited by SPEED_MIN and SPEED_MAX. Within this speed range, controller 16 selects control values that ensure that the constant average power is
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51/57 reached between SPEED_CP_MIN and SPEED_CP_MAX. Controller 16 also drives motor 2 over a voltage range limited by V_MIN and V_MAX. Within this voltage range, controller 16 selects control values that ensure that the average constant power is achieved between V_CP_MIN and V_CP_MAX.
Controller 16 stores three forward lookup tables. The first feed lookup table is a one-dimensional table that is indexed by the speed of the rotor when operating in multi-switching mode. The second lookup table is a one-dimensional table that is indexed by the speed of the rotor when operating in single overcurrent switching mode. The third lookup table is a two-dimensional table that is indexed by the speed of the rotor and the voltage when operating in the single switching mode of unlimited free rotation.
The free spin lookup table, timeout lookup table and advance lookup table used in multi-switching mode are stored collectively by controller 16 as a single multi-switching map. Figure 24 details the multi-switching map used by controller 16. The map stores a free spin period, T_FW, a timeout period, T_TO and an advance period, T_ADV, for each of a plurality of speeds. The electrical angles corresponding to the various periods are also listed. However, the angles are not part of the map stored by the controller 16 and are provided merely to illustrate the behavior of the angles with the speed of the rotor. For example, it can be seen that a fixed timeout period, T_TO, of 70 ps is used throughout the Acceleration at Average Speed. However, the corresponding timeout angle, A_TO, increases from 8.4 degrees at 10 krpm to 42.0 degrees at 55 krpm.
The advance and overheat lookup tables used in the single overcurrent switching mode are stored in a similar way as a single map. A part of the map is detailed in Figure 25. Again, the corresponding electrical angles are provided for illustration purposes.
The feed, lead offset, and lead phase shift tables used in single unlimited free-spin switching mode are also stored as a single map. This means that the same speed and voltage resolution are used for each of the three look-up tables. Consequently, each element of the map stores a lead time, a lead offset value and a lead phase offset value. However, for the sake of clarity, a part of each lookup table is illustrated in Figures 26-28; the unit for all three lookup tables is ps. Instead of storing absolute values, the advance lookup table and the drift lookup table store difference values. Controller 16 then stores a reference lead time of 56.2 ps and a lead deviation value of 870190080086, from 19/08/2019, pg. 58/93
52/57 reference voltage of 48.8 ps, corresponding to a speed of 94 krpm and an RMS voltage of 230 V.
Figure 29 details a section of the driving sine lookup table that is stored and used by controller 16 in the unlimited free-spin single switching mode. Since the frequency of the AC supply is 50 Hz, the lookup table spans from 0 to 0.01 second, which corresponds to an AC power semiconductor 4. The lookup table resolution is 51.2 pseg, and the conduction amplitude, T_CD_AMP, and the conduction phase angle, T_CD_PHASE are 83.2 ps and 320 ps, respectively. The driving phase angle, T_CD_PHASE, is effectively a reference phase shift for a speed of 94 krpm and a voltage of 230 V.
Motor system 1 has an operating speed range of 81 krpm to 106 krpm and an operating voltage range of 200 V to 260 V. Within these ranges, an average input power of 1600 W ± 25 W is maintained at speeds between 85 krpm and 106 krpm and at voltages between 219 V to 256 V. Furthermore, an efficiency of about 85% is achieved over the voltage range and constant power speed.
Controller 16 is a PIC16F690 microcontroller manufactured by Microchip Technology Inc. This is a simple 8-bit microcontroller with a clock speed of 20 MHz, a single ADC, two comparators, three timers, 4096 words of program memory and 512 bytes of data memory. Even with this relatively simple microcontroller, controller 16 is capable of driving motor 2 at speeds in excess of 100 krpm, with an average input power of around 1600 W.
Until then, reference was made to a driving period having a waveform that varies like a semisinoid during each cycle of the waveform (and therefore during each AC 4 semiconductor cycle). However, other types of periodic waveforms can be used for the driving period. In particular, it has been found that waveforms for which the conduction period varies like a triangle or trapezoid during each cycle of the waveform work well to obtain a relatively satisfactory power factor. Figure 30 illustrates the voltage waveform of the AC 4 supply along with the three waveforms mentioned above for the conduction period: (a) semisinoid; (b) triangle; and (c) trapezoid. For each of these waveforms, the driving period increases during the first half of each waveform cycle and decreases during the second half of the cycle. Of these three waveforms, it was found that the semisenoid offers the best results in terms of lower order harmonics. However, for an engine system with different characteristics, it is reasonably possible that the improved performance can be achieved using a different waveform.
In the embodiments described above, the advance period is constant during each half cycle of the AC power 4. This then simplifies the instructions executed by the ControPetição 870190080086, of 19/08/2019, p. 59/93
53/57 lador 16. However, it is possible to achieve greater performance by employing an advance period that varies throughout each AC power semiconductor 4.
When switching the winding in phase 7 before the edges of the HALL signal, the DC link voltage used to excite the winding in phase 7 is intensified by the force against the electromotive. As a result, the direction of the current through the phase 7 winding can be reversed more quickly. In addition, the current in the winding in phase 7 can be made to conduct the force against the electromotive, so that more current can be drawn into the winding in phase 7 during the period of positive torque. As the DC link voltage increases, the time required to reverse the direction of the phase current decreases and the speed at which the phase current rises increases. Therefore, a shorter advance period can be used and any deficit in the amount of current in phase can be compensated by increasing the conduction period. It is important to note that, by decreasing the advance period, the negative torque period is reduced and, therefore, a more efficient engine system 1 can be realized. Controller 16 can therefore employ a lead time that varies over each AC power semiconductor 4. For that purpose, the lead time duration can be defined by a periodic waveform, each cycle of the waveform repeating with each AC 4 semiconductor cycle. The advance period is longer in the region close to zero crossings in the AC 4 supply voltage, and is shorter in the region near the peak voltage. Waveforms suitable for the advance period include inverted semisinoid, inverted triangle and inverted trapezoid. Figure 31 illustrates the voltage waveform of the AC 4 supply along with the three possible waveforms for the advance period: (a) inverted semisinoid; (b) inverted triangle; and (c) inverted trapezoid. The lead time is defined, stored and updated by controller 16 in much the same way as described above for the lead time. For example, a lead time, T_ADV, which varies as an inverted semisinoid, can be defined as:
T_ADV = T_ADV_OFFSET + T_ADV_AMP * abs {sen ({t * f * 360 degrees)}
Controller 16 then uses the time that has elapsed since a zero crossover in the AC supply voltage 4 to determine a lead time for each electric semicycle of motor 2. Controller 16 can additionally update the lead time waveform in response changes in the rotor speed and / or changes in the RMS voltage of the AC supply 4. For example, controller 16 can adjust one or more of the deviation, amplitude and phase of the waveform in response to changes in the speed of the rotor and / or tension. Again, just like the driving period, the lead time can be defined as the sum of two components: a first component that is constant and a second component that varies over each cycle of the lead period waveform. Controller 16 then adjusts one or both components to
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54/57 response to changes in rotor speed and / or RMS voltage.
The parameters (eg lead time, drift, etc.) used by controller 16 are adjusted in response to changes in the RMS voltage of the AC supply 4 only when operating in Travel Mode. This then reduces the size of the lookup tables that are used by controller 16 during acceleration. As a result, more memory is made available for the most important look-up tables used during Travel Mode. However, there may be cases where it is desirable to adjust one or more parameters during acceleration in response to changes in the RMS voltage of the AC supply 4. For example, without adjusting the control values, motor system 1 can start with power greater or less if the RMS voltage of the AC 4 supply is greater or less than a nominal voltage. By adjusting the parameters during acceleration, better control over power can be achieved. Controller 16 can therefore store a voltage compensation table for one or more of the parameters used during acceleration, for example, free spin period, timeout period, advance period and overheat period. The voltage compensation table stores a compensation value for each of a plurality of voltages. When updating a particular parameter, controller 16 indexes the relevant lookup table using the rotor speed to select a control value. In addition, controller 16 indexes the relevant voltage compensation table using the RMS voltage of the AC supply 4 to select a compensation value. Controller 16 then adds the control value and the voltage compensation value to obtain the parameter value. In this particular example, the voltage compensation table is one-dimensional. However, any voltage compensation preferably depends not only on the RMS voltage of the AC supply 4, but also on the speed of the rotor 5. Therefore, instead of storing two one-dimensional look-up tables for each parameter, controller 16 can , instead, store a complete two-dimensional lookup table for each parameter, as is done for the parameters used during Travel Mode (for example, Figure 26-28). However, a complete two-dimensional table requires considerably more memory than two one-dimensional tables.
In the embodiments described above, the motor 2 comprises a four-pole rotor 5 and a four-pole stator 6. However, the rotor 5 and stator 6 could have a smaller or greater number of poles. As the number of poles increases, the number of electrical cycles per mechanical cycle increases. Consequently, for a given rotor speed, each Hall period is shorter. A faster controller 16 could therefore be needed in order to carry out the necessary instructions during each Hall period. In addition, faster switching to inverter 10 could be necessary. As a result, the number of permissible poles tends to be limited by the operating speed of motor 2 and / or
Petition 870190080086, of 8/19/2019, p. 61/93
55/57 by the control system components 3.
The current controller 22 described above and illustrated in Figure 5 makes use of the internal peripherals of the PIC16F690 microcontroller. Alternative configurations for current controller 22 are possible, depending on the particular microcontroller that is used for controller 16. Furthermore, it is not essential that current regulator 22 forms part of controller 16. Instead, the current regulator 22 can be formed separately from controller 16. Controller 16 would then include an input pin 20 coupled to current regulator 22 to receive the overcurrent signal.
The position sensor 13 used by motor system 1 is a Hall effect sensor. However, alternative position sensors capable of generating a signal indicating the position of the rotor 5 could also be employed, for example, an optical sensor. Similarly, instead of employing a pair of fixing diodes, other configurations can be used for the zero crossing detector 12, for example, the Schmitt trigger.
Controller 16 freely rotates the winding in phase 7 by opening the switches on the high side Q1, Q2 of the inverter 10. This then allows the current in the winding in phase 7 to recirculate around the loop on the lower side of the inverter 10. It is conceivable allow the free rotation to occur by opening the switches on the lower side Q3, Q4 and allowing the current to recirculate around the loop on the upper side of the inverter 10. However, it would then be necessary for the derivative resistors R1, R2 of the current 12 are located in the upper arms of the inverter 10 so that the current can continue to be detected during free rotation. This, in turn, would lead to greater power losses, since the resistors in derivation R1, R2 would be subject to higher voltages during excitation. In addition, the voltage across the shunt resistors R1, R2 would be fluctuating instead of being referenced to neutral, and thus measuring the current in the phase 7 winding would be difficult.
In the embodiments described above, motor system 1 comprises a control system 3 that drives a permanent magnet motor 2. However, many aspects of control system 3 could also be used to drive other types of brushless motors.
The use of a driving period and / or an advance period that varies from time to time can be used to excite the phase windings of other types of brushless motors, for example, reluctance motors. For a reluctance motor, the rotor does not include an electromotive force in the motor phase windings. Therefore, it is possible to obtain a substantially sinusoidal current waveform without the need for a variable conduction period or lead time. However, a lead time and / or lead time that varies during each AC 4 power cycle can be used to obtain a particular envelope for the flow density.
Petition 870190080086, of 8/19/2019, p. 62/93
56/57 magnetic on the engine.
Thanks to the ripple in the DC link voltage, the windings of the reluctance motor are excited with a voltage that varies during each AC 4-power cycle. If a constant driving period was used during each AC 4-power cycle, the density envelope of magnetic flux in the motor would reflect that of the DC link voltage. The controller 16 can therefore employ a driving period which varies during each AC power semiconductor 4 in order to conform the magnetic flux density envelope. In particular, controller 16 may employ a driving period that reduces the peak magnetic flux density. By reducing the peak magnetic flux density, it is possible to make a motor more efficient and / or smaller. In order to reduce the peak magnetic flux density, controller 16 employs a conduction period that is longer in the region around the zero crossings in the AC 4 supply voltage, and is shorter in the region around the peak voltage. . Waveforms suitable for the driving period include semisinoid, inverted triangle and inverted trapezoid.
In order to compensate for the variation in the length of the driving period, the controller 16 can additionally employ an advance period that varies periodically with time. In particular, as the driving time decreases, the controller 16 can employ a longer lead time to compensate for the shorter driving time. Therefore, unlike the driving period, controller 16 employs a lead time that is short in the region around the zero crossings in the AC supply voltage 4, and which is longer in the region around the peak voltage. Waveforms suitable for the advance period include semisinoid, triangle and trapezoid.
For the permanent magnet motor 2, the controller 16 excites the winding in phase 7 before the zero crossings of the electromotive force in the winding in phase 7, for example, as determined by the signal generated by the position sensor 13. For a motor reluctantly, controller 16 excites the winding before increasing the inductance, which can again be determined by means of a position sensor. In both cases, the controller 16 excites the phase winding before the predetermined positions of the rotor. More particularly, the controller 16 excites the phase winding before the non-aligned positions of the rotor.
Although a driving period and / or a lead period that varies periodically over time can be used with different types of brushless motors, variable lead and / or lead times are of particular benefit when used to drive a magnet motor. permanent. As noted above, the counter electromotive force induced in the phase 7 winding by the permanent magnet rotor 5 makes it difficult to precisely control the amount of current drawn from the AC supply.
Petition 870190080086, of 8/19/2019, p. 63/93
57/57 of a conduction period that varies periodically with time, a waveform that approximates that of a sine wave can be obtained for the current drawn from the AC power source without the need for an active PFC connection capacitor or high capacitance.
Updating the control parameters (for example, lead time, lead time, free spin period and timeout period) in response to zero crossings in the voltage of an AC supply can be used with other types of brushless motors . As noted above, by updating a control parameter in response to zero crossings on the AC supply, the control parameter is updated at regular intervals, regardless of the motor speed. In addition, the control parameter is updated regularly without the need for a dedicated timer. The control parameter is also updated in sync with the AC power cycle. As a result, the current waveform drawn from AC power is generally more stable.
Interruption conflict is a possible problem for many types of brushless motors. Consequently, the use of a timer and comparator (for example, forming part of a dedicated comparator module or as part of a PWM module) to generate a control signal in hardware instead of software can be used with other types of brushless motors in order to reduce the total number of interruptions. In addition, when it is necessary for the controller to sample an analog signal, the interruption conflict can be further reduced by dividing the sampling process into a series of steps, each step being performed in response to a different edge of the sensor signal. position. As a result, the sampling process is spread over a series of electric motor cycles, thereby freeing up more time for the controller to perform other routines during each electrical cycle.

权利要求:
Claims (19)
[1]
1. Method of controlling a brushless permanent magnet motor (2), the method comprising:
rectify an alternating voltage to provide a rectified voltage with a ripple of at least 50%; and excite a winding (7) of the motor (2) with the voltage rectified, the winding (7) being excited before the zero crossings of the counter electromotive force for a period of advance and being excited for a period of driving throughout each electric motor semicycle (2),
CHARACTERIZED by the fact that the method also comprises adjusting one of the advance period and the driving period in response to changes in one of the motor speed and the RMS value of the alternating voltage in order to keep the average power constant.
[2]
2. Method, according to claim 1, CHARACTERIZED by the fact that the method comprises adjusting one of the advance period and the driving period in order to maintain a power factor of at least 0.95.
[3]
3. Method, according to claim 1 or 2, CHARACTERIZED by the fact that the method comprises adjusting one of the advance period and the driving period in order to maintain an efficiency of at least 80%.
[4]
4. Method according to one of claims 1 to 3, CHARACTERIZED by the fact that the alternating voltage has an RMS value between 100 V and 240 V and the method comprises adjusting one of the advance period and the driving period so to maintain a constant average power of at least 1000 W.
[5]
5. Method according to any one of claims 1 to 4, CHARACTERIZED by the fact that the method comprises adjusting one of the advance period and the driving period in order to maintain the constant average power over one of a range speed that covers 10 krpm and a voltage range that covers 10 V.
[6]
6. Method, according to claim 5, CHARACTERIZED by the fact that the speed range has a minimum value greater than 60 krpm and a maximum value greater than 80 krpm.
[7]
7. Method, according to claim 5 or 6, CHARACTERIZED by the fact that the speed range has a maximum value greater than 100 krpm.
[8]
8. Method, according to any one of claims 1 to 7, CHARACTERIZED by the fact that the duration of one of the advance period and the driving period is defined by a waveform that varies along each voltage semiconductor alternating, and the method comprises adjusting the waveform in response to changes
Petition 870190080086, of 8/19/2019, p. 65/93
2/3 tions in one of the motor speed and the RMS value.
[9]
9. Method, according to claim 8, CHARACTERIZED by the fact that the duration of one of the advance period and the driving period comprises the sum of a first component and a second component, the first component being constant during each semicycle alternating voltage, the second component varying during each alternating voltage semicycle, and the method comprises adjusting the first component in response to a change in one of the motor speed and the RMS value.
[10]
10. Method, according to claim 9, CHARACTERIZED by the fact that the second component is defined by the length of time that has elapsed since a zero crossing at the alternating voltage.
[11]
11. Method according to any one of claims 1 to 10, CHARACTERIZED by the fact that the method comprises adjusting one of the advance period and the conduction period in response to a zero crossing in the alternating voltage.
[12]
12. Method, according to claim 11, CHARACTERIZED by the fact that the advance period is constant during each alternating voltage semicycle, and the method comprises adjusting the advance period in response to crossing at zero.
[13]
13. Method, according to claim 11 or 12, CHARACTERIZED by the fact that the method comprises storing a first lookup table of first control values, indexing the first lookup table in response to crossing at zero, the first lookup table query being indexed using one of the speed and voltage to select a first control value, and use the first control value to determine the lead time.
[14]
14. Method according to any one of claims 11 to 13, CHARACTERIZED by the fact that the duration of the driving period comprises the sum of a first component and a second component, the first component being constant and the second component varying during each alternating voltage semicycle, and the method comprises storing a second lookup table for second control values, indexing the second lookup table in response to crossing at zero, the second lookup table being indexed using one of the speed and voltage for select a second control value, and use the second control value to determine the first component.
[15]
15. Control system (3) for a brushless permanent magnet motor (2), CHARACTERIZED by the fact that the control system (3) performs a method as defined in any of claims 1 to 14.
[16]
16. Control system, according to claim 15, CHARACTERIZED by
Petition 870190080086, of 8/19/2019, p. 66/93
3/3 the control system (3) comprises:
a rectifier (8) to rectify the alternating voltage;
an inverter (10) coupled to the winding (7); and a controller (16) to control the inverter (10), where the controller (16) generates one or more control signals to excite the winding (7) before zero crossings against electromotive force (EMF), the inverter (10), in response to the control signals, excites the winding (7) with the rectified voltage, and the controller (16) adjusts one of the advance period and the driving period in response to changes in one of the speed of the motor and the RMS value.
[17]
17. Control system, according to claim 16, CHARACTERIZED by the fact that the control system (3) comprises a position sensor (13) that generates a signal having edges that correspond to zero crossings of the force against electromotive ( EMF), and the controller (16) generates the control signals before each edge of the signal.
[18]
18. Control system according to claim 16 or 17, CHARACTERIZED by the fact that the control system (3) comprises a zero crossover detector (14) to detect zero crossings at alternating voltage, and the controller (16) sets one of the lead time and lead time in response to a zero crossing at the alternating voltage.
[19]
19. Motor system (1), CHARACTERIZED by comprising a brushless permanent magnet motor (2) and a control system (3) as defined in any of claims 15 to 18.

类似技术:

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公开号 | 公开日 | 专利标题

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BR112012026387B1|2019-11-12|Method and control system of a brushless permanent magnet motor, and motor system

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同族专利:

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公开号 | 公开日

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KR20130027516A|2013-03-15|

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CN102939713A|2013-02-20|

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BR112012026387A2|2016-08-02|

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GB2480729B|2013-03-20|

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GB201106305D0|2011-05-25|

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GB2480729A|2011-11-30|

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EP2559158A2|2013-02-20|

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WO2011128686A2|2011-10-20|

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US8937446B2|2015-01-20|

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AU2011239773A1|2012-11-01|

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EP2559158B1|2018-11-14|

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KR20140057381A|2014-05-12|

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JP2011229388A|2011-11-10|

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JP5546496B2|2014-07-09|

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MX2012012053A|2013-03-15|

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KR101482553B1|2015-01-14|

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GB201006391D0|2010-06-02|

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CN102939713B|2015-06-03|

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AU2011239773B2|2014-05-29|

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US20110254488A1|2011-10-20|

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GB201105209D0|2011-05-11|

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WO2011128686A3|2012-12-20|

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法律状态:
2018-12-26| B06F| Objections, documents and/or translations needed after an examination request according [chapter 6.6 patent gazette]|

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2019-06-18| B06T| Formal requirements before examination [chapter 6.20 patent gazette]|

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2019-10-22| B09A| Decision: intention to grant [chapter 9.1 patent gazette]|

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2019-11-12| B16A| Patent or certificate of addition of invention granted|Free format text: PRAZO DE VALIDADE: 20 (VINTE) ANOS CONTADOS A PARTIR DE 13/04/2011, OBSERVADAS AS CONDICOES LEGAIS. (CO) 20 (VINTE) ANOS CONTADOS A PARTIR DE 13/04/2011, OBSERVADAS AS CONDICOES LEGAIS |

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2021-04-06| B21F| Lapse acc. art. 78, item iv - on non-payment of the annual fees in time|Free format text: REFERENTE A 10A ANUIDADE. |

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2021-08-10| B24J| Lapse because of non-payment of annual fees (definitively: art 78 iv lpi, resolution 113/2013 art. 12)|Free format text: EM VIRTUDE DA EXTINCAO PUBLICADA NA RPI 2622 DE 06-04-2021 E CONSIDERANDO AUSENCIA DE MANIFESTACAO DENTRO DOS PRAZOS LEGAIS, INFORMO QUE CABE SER MANTIDA A EXTINCAO DA PATENTE E SEUS CERTIFICADOS, CONFORME O DISPOSTO NO ARTIGO 12, DA RESOLUCAO 113/2013. |

优先权:

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申请号 | 申请日 | 专利标题

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GBGB1006391.5A|GB201006391D0|2010-04-16|2010-04-16|Control of a brushless permanent-magnet motor|

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PCT/GB2011/050733|WO2011128686A2|2010-04-16|2011-04-13|Control of a brushless permanent-magnet motor|

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