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    • 专利摘要:
    WATERPROOF SIGNAL PROVIDER AND METHOD TO PROVIDE A WATERPROOF SIGN. A watermark signal provider for providing a watermark signal depending on a frequency and time domain representation of watermark data, where the frequency and time domain representation comprises associated values At frequency sub-bands and bit intervals, the watermark signal provider comprises a frequency and time domain waveform provider to provide time domain waveforms for a plurality of frequency and time sub-bands. watermark data. The frequency and time domain waveform provider is configured to map a given value of the frequency and time domain representation into a bit-forming function. A time extension of the bit-forming function is greater than the bit interval associated with the given value of the frequency and time domain representation, so that there is a temporal overlap between functions formed by bits provided for temporally subsequent values of the domain representation in (...).
    公开号:BR112012021533B1
    申请号:R112012021533-7
    申请日:2011-02-23
    公开日:2020-11-10
    发明作者:Reinhard Zitzmann；Stefan WABNIK；Joerg Pickel；Bert Greevenbosch；Ernst Eberlein；Bernhard Grill；Stefan Krägeloh；Giovanni Del Galdo；Tobias Bliem；Juliane Borsum；Marco Breiling
    申请人:Fraunhofer-Gesellschaft zur Förderung der angewandten Forschung e.V.；
    IPC主号:
    专利说明:

    DESCRIPTION TECHNICAL FIELD
    The embodiments according to the present invention relate to a watermark signal provider to provide a watermark signal depending on a representation of the watermark data frequency and time domain. Other achievements refer to a method for providing a watermark signal depending on a frequency and time representation of watermark data.
    Some realizations, according to the invention, refer to a system of creation of watermark of resistant audio, of low complexity. HISTORY OF THE INVENTION
    In many technical applications, it is desired to include extra information in an information or signal that represents useful data or "main data", such as, for example, an audio signal, a video signal, graphics, a measurement quantity and so on. against.
    In many cases, it is desired to include the extra information, so that the extra information is linked to the main data (for example, audio data, video data, still image data, measurement data, text data and so on) ), so that it is not perceived by a user of said data.
    Also, in some cases, it is desirable to include the extra data, so that the extra data is not easily removable from the main data (for example, audio data, video data, still image data, measurement data and so on. ).
    This is particularly true in applications where it is desirable to implement digital rights management4.
    However, sometimes it is simply desired to add substantially imperceptible parallel information to the useful data.
    For example, in some cases, it is desirable to add parallel information to audio data, so that the parallel information provides information about the source of the audio data, the content of the audio data, rights related to the audio data and so on.
    To embed extra data in the useful data or "main data", a concept called "watermark creation" can be used.
    The concepts of watermark creation have been discussed in the literature for many different types of useful data, such as audio data, still image data, video data, text data and so on.
    Below, some references will be given, in which the concepts of watermark creation are discussed.
    However, the reader's attention is also directed to the wide range of literature in books and publications related to the creation of the watermark for further details.
    DE 196 40 814 C2 describes an encryption method for introducing an non-audible data signal into an audio signal and a method for decoding a data signal, which is included in an audio signal in an audible form.
    The encoding method for introducing a non-audible data signal into an audio signal comprises converting the audio signal into the spectral domain.
    The encoding method also comprises determining the masking threshold of the audio signal and the provision of a pseudo-noise signal. The encoding method also comprises providing the data signal and multiplying the pseudo noise signal with the data signal in order to obtain a propagated frequency data signal.
    The encoding method further comprises the weighting of data signal propagation with the masking and overlap limit of the audio signal and the weighted data signal. In addition, WO 93/07689 describes a method and equipment for automatically identifying a program broadcast by a radio station or television channel or recorded on a media by adding an inaudible encoded message to the program's sound signal, the message identifying the channel or broadcast station, the program and / or the exact date .
    In an embodiment discussed in said document, the sound signal is transmitted via an analog to digital converter to a data processor that allows the frequency components to be separated and that allows the energy in some frequency components to be changed a predetermined way to form a coded identification message. The output of the data processor is connected by a digital to analog converter to an audio output for transmission or recording of the sound signal.
    In another embodiment discussed in said document, an analog pass band is used to separate a frequency band from the sound signal, so that the energy in the separate band can thus be changed to encode the sound signal.
    US 5,450,490 describes equipment and methods for including a code that has at least one component of * code frequency in an audio signal. The capabilities of the various frequency components in the audio signal to mask the * code frequency component to human hearing are assessed and, based on these assessments, a code frequency component amplitude is assigned.
    The methods and equipment for detecting a code in an encoded audio signal are also described. A code frequency component in the encoded audio signal is detected based on an expected code amplitude or a noise amplitude within a range of audio frequencies, including the frequency of the code component.
    WO 94/11989 describes a method and equipment for encoding / decoding broadcast or recorded segments and monitoring audience exposure to it. Methods and equipment for encoding and decoding information on broadcast or recorded segment signals are described.
    In an implementation described in the document, an audience monitoring system encodes the identification information in the audio signal part of a transmission or recorded segment using propagation spectrum encryption. The monitoring device receives an acoustically reproduced version of the transmission signal or recorded using a microphone, decodes the identification information of the audio signal part, despite the significant environment, and stores this information, automatically providing a diary for the audience member, which is then upgraded to a centralized installation.
    A separate monitoring device decodes the additional information from the transmission signal, which is matched with the * audience journal information at the central facility. This monitor can simultaneously send data to the centralized unit, using a dial-up telephone line, and receives data from the centralized unit via an encrypted signal, using a spread spectrum technique and modulated with a third party transmission signal.
    WO 95/27349 describes equipment and methods for including codes in audio and decoding signals. An apparatus and methods for including a code having at least one code frequency component in an audio signal are described.
    The capabilities of various frequency components in the audio signal to mask the code frequency component to human hearing are assessed and, based on these assessments, an amplitude is assigned to the code frequency components. Methods and equipment for detecting a code in an encoded audio signal are also described.
    A code frequency component in the encoded audio signal is detected based on the expected code amplitude or a noise amplitude within a range of audio frequencies, including the frequency of the code component. watermark creation, a watermark signal is based on a plurality of adjacent time-domain waveforms, in which a maximum energy of these waveforms is limited, since the d 'signal water must be kept inaudible.
    However, a low energy of the waveform and, therefore, of the watermark signal, leads to a more difficult detection of the watermark signal and can lead to bit errors and, therefore, a low resistance of * watermark signal. In view of this situation, it is the objective of the present invention to create a concept to provide a watermark signal that allows easier decoding of the watermark signal on the side of a receiver. SUMMARY OF THE INVENTION
    The objective is achieved by a watermark signal provider according to claim 1, a method for providing a watermark signal according to claim 10 and a computer program according to claim 11. An achievement of According to the present invention, it creates a watermark signal provider to provide a watermark signal depending on a representation of frequency and time domain watermark data.
    The representation of the frequency and time domain comprises values associated with the frequency bands and bit intervals.
    The watermark signal provider comprises a frequency and time domain waveform provider and a time domain waveform combiner.
    The time domain frequency waveform provider is configured to map a given frequency and time domain representation value into a bit deformation function. A time extension of the debits formation function is greater than the bit interval associated with the determined value of the representation of the frequency and time domain, so there is a time overlap between functions formed by bits provided for temporally subsequent values of the representation of the frequency and time domain. same frequency sub-range.
    The frequency and time domain waveform provider is
    * still configured so that a domain waveform within a given frequency sub-range contains one, plurality of functions formed by bits provided for the time-subsequent value of the frequency and time domain representation of the same frequency range. The time domain waveform combiner is configured to combine the waveforms provided for the plurality of frequencies of the frequency and time domain waveform provider to derive the watermark signal.
    The main idea of the present invention is not only to correlate binary values (that is, binary values of the same frequency sub-range and subsequent bit intervals) of a watermark data representation, but also to correlate the functions formed by bits corresponding to these values with each other . In this way, a redundancy in the watermark signal is added, which allows easier decoding on the side of a receiver, without increasing the energy of the watermark signal.
    In addition, the resistance of the watermark signal is greater. This correlation of the bit-formed function is obtained in the realizations by the bit-forming function, in which a time extension of the bit-forming function is greater than a bit time of corresponding values. representation of frequency and time domain.
    Therefore, a decoder for the watermark signal on the side of a receiver can be made easier and less complex than a decoder for a conventional watermark creation system. In addition, a chance of obtaining correct watermark information from a signal obtained may be greater, especially in noisy environments.
    The values of the frequency domain and time representation of watermark data can be binary values, where a value corresponds to a frequency sub-range and a bit range.
    In one embodiment, the frequency and time domain waveform provider is configured to provide a bit-formed function for each of the frequency and time domain representation values, where the frequency and time domain name provider is configured so that functions formed by bits of adjacent values in the same frequency range overlap and therefore a correlation of functions formed by bits of adjacent values is achieved.
    In one embodiment, the frequency and time domain waveform provider can be configured so that a function formed by bits provided for a given value of the representation of frequency and time domain is superimposed by a function formed by bits of a value temporally previous to the same frequency sub-range as the determined value of the frequency and time domain representation value and by a bit-formed function of a value temporally following the same frequency range as the determined value of the frequency and time domain representation, so that a domain time provided by the frequency domain waveform provider and time contain an overlap between at least three functions formed by temporally subsequent bits of the same frequency range. In other words, a time domain waveform for a given frequency sub-band is in a given bit range at least based on a „first function formed by bits of a first value corresponding to the given frequency sub-band and a given time interval, in a second function formed by bits of a second value corresponding to the determined frequency sub-range and a time interval earlier in time; and in a third function formed by bits of a third value corresponding to the determined frequency sub-range and a time interval following time. The time extension of a bit-forming function can be a time range, in which the bit-forming function comprises values other than zero.
    In addition, the time range, in which the bit-forming function comprises non-zero values, can be at least three bit intervals in length. A bit-forming function can also be called a bit-creating function and can be different for each frequency sub-range of the frequency and time representation of the watermark data.
    In this way, therefore, a different filtering (bit formation) is obtained for different frequency sub-bands.
    In one embodiment, a bit-forming function can be based on an amplitude-modulated periodic signal. An amplitude modulation of the amplitude-modulated periodic signal can be based on a base band function.
    A time extension of the bit-forming function can be based on the baseband function. Therefore, a time span of the baseband function, where the baseband function contains values other than zero, is greater than the bit range.
    The function of> base range can be identical to values in the same range of frequency representation of the frequency and time domain of the watermark data.
    In one embodiment, the baseband function is identical for a plurality or all frequency sub-bands of the frequency and time domain representation. In other words, the base range function can be the same for a plurality of values or all values of the representation of the frequency and time domain.
    If the non-identical baseband function for each sub-range, more efficient implementation on the side of a decoder is possible. In one embodiment, a modulation factor for the breadth of a bit-forming function may be a baseline time domain function. , for example, as a filter function.
    The base band function can be identical for values of the same frequency range of the frequency and time domain representation of the watermark data. In one embodiment, a periodic part of a bit-forming function of a given frequency sub-band can be based on a cosine function, based on a frequency that is a central frequency of the given frequency sub-range.
    In one embodiment, the watermark signal provider further comprises a weight tuner, for example, a psychoacoustic processing module, which is configured to tune a weight (and therefore an amplitude) of each function formed by bits for each value of the time domain representation of watermark data. The weight tuner can be »configured to maximize a function's energy formed by bits of a certain value in relation to the inaudibility of the watermark signal.
    In other words, the weight tuner can be configured to fine tune the weights to assign the maximum possible energy to the watermark, while keeping it audible. In one embodiment, the weight tuner can be configured to tune the weights in an iterative process controlled by weight tuner.
    The weight tuner can therefore adjust each bit-formed function provided from the frequency and time domain waveform provider, so that each bit-formed function has maximum energy (but, of course, remains inaudible) and, therefore, it is better to detect on the side of a decoder. In one embodiment, a time domain waveform for a given frequency sub-band is a sum of all the bit-formed functions of the given frequency sub-band.
    In one embodiment, the watermark signal is a sum of the waveforms provided for the plurality of frequency sub-bands. Some embodiments in accordance with the invention also create a method for providing a watermark signal depending on a domain representation frequency and time of watermark data.
    This method is based on the same findings as the equipment discussed earlier.
    Some embodiments according to the invention comprise a computer program for carrying out the methods of the ► invention. BRIEF DESCRIPTION OF THE FIGURES
    The realizations, according to the invention, will be subsequently described, with reference to the attached figures, in which: Figure ■ 1 presents a schematic block diagram of a watermark inserter, according to an embodiment of the invention; Figure 2 presents a schematic block diagram of a watermark decoder, according to an embodiment of the invention; Figure 3 presents a detailed schematic block diagram of a watermark generator, according to an embodiment of the invention; Figure 4 presents a detailed schematic block diagram of a modulator for use in carrying out the invention.
    Figure 5 shows a detailed schematic block diagram of a psychoacoustic processing module for use in an embodiment of the invention; Figure 6 shows a schematic block diagram of a psychoacoustic model processor for use in an embodiment of the invention; Figure 7 is a graphical representation an energy spectrum of an audio signal produced by block 801 at frequency; Figure 8 is a graphical representation of an energy spectrum of an audio signal produced by block 802 • at frequency; Figure 9 shows a schematic block diagram of an amplitude calculation; Figure 10a shows a schematic block diagram of a modulator; Figure 10b is a graphical representation of the location of coefficients in the frequency claim and time; Figures 11a and 11b show schematic block diagrams of alternative implementations of the de-synchronization module; Figure 12a is a graphic representation of the problem in finding the time alignment of a d 'mark. water; Figure 12b is a graphical representation of the problem at the beginning of the message identification; Figure 12c is a graphical representation of a timeline of synchronization sequences in a complete message synchronization mode; Figure 12d is a graphical representation of the time alignment of the sequences synchronization in a partial message synchronization mode; Figure 12e is a graphical representation of the synchronization module input data; Figure 12f is a graphic representation of a synchronization beat identification concept; Figure 12g shows a schematic block diagram of a synchronization signature correlator; Figure 13a is a graphical representation of a, example for a temporal de-propagation; Figure 13b is a graphic representation of an example for a multiplication by element between the debits and propagation sequences; Figure 13c is a graphic representation of an output of the signature correlator synchronization, after temporal measurement; Figure 13d is a graphical representation of a filtered synchronization signature correlator output with the auto-correlation function of the synchronization signature; Figure 14 shows a schematic block diagram of a watermark extractor, according to an embodiment of the invention; Figure 15 is a schematic representation of a selection of a part and the representation of frequency and time domain as a candidate message; Figure 16 shows a schematic block diagram of an analysis module; Figure 17a is a graphical representation of an output of a synchronization correlator; Figure 17b is a graphical representation of messages decoded; Figure 17c is a graphical representation of a synchronization position, which is extracted from a watermarked signal; Figure 18a is a graphical representation of a payload, a payload with a Viterbi closure sequence, a payload coded by Viterbi and a version.
    encoded by repetition of the payload encoded by Viterbi; Figure 18b is a graphic representation of unloaders used to embed a watermarked signal; Figure 19 is a graphic representation of an unencrypted message, a coded message, an out of sync message and a watermark signal, in which the synchronization sequence is applied to messages; Figure 20 is a schematic representation of a first stage of a so-called "ABC synchronization" concept; Figure 21 is a graphical representation of a second stage of a so-called concept "ABC synchronization"; Figure 22 is a graphical representation of a third stage of a so-called "ABC synchronization" concept; Figure 23 is a graphic representation of a message comprising a payload and a CRC portion; Figure 24 presents a diagram of schematic blocks of a watermark signal provider, according to an embodiment of the invention; and Figure 25 shows a flowchart of one m This is to provide a watermark signal depending on a representation of the frequency and time domain, according to an embodiment of the invention. DETAILED DESCRIPTION OF ACHIEVEMENTS 1. WATERPROOF SIGNAL PROVIDER
    In the following, a watermark signal provider 2400 will be described, referring to Figure 24, which shows a schematic block diagram of said watermark signal provider.
    The 2400 watermark signal provider is configured to receive watermark data, such as a 2410 frequency time domain representation on an input and to provide, based on this, a 2420 watermark signal on an output.
    The 2400 watermark generator comprises a 2430 frequency and time domain waveform provider and a 2460 time domain waveform combiner.
    The 2430 frequency and time domain waveform provider is configured to provide 2440 time domain waveforms for a plurality of frequency sub-bands, based on the 2420 frequency and time domain representation of the watermark data.
    The 2430 frequency and time domain waveform provider is configured to map a given 2410 frequency and time domain representation value into a 2450 bit-forming function.
    A time extension of the bit deformation function 2450 is greater than the bit interval associated with a given value of the frequency domain representation and time 2410, so that there is a temporal overlap between bit-formed functions provided for temporally subsequent values of the frequency domain representation and time2410 of the same frequency sub-range.
    The provider of the frequency domain and time domain 2430 is further configured to modify the time domain waveform 24 4 0 of a given frequency band containing a plurality of functions formed by bits provided for time-subsequent values of the domain representation of frequency and time 2410 of the same frequency range.
    The 2460 time domain waveform combiner is configured to combine the 2440 provided waveforms for the frequency range 2430 frequency and time domain waveform provider to derive the 2420 watermark signal.
    According to one embodiment, the 2430 frequency and time domain waveform provider may comprise a plurality of bit-forming blocks configured to map a given value of the frequency domain representation and time 2410 of the watermark data into a function For formation bits 2450, the outputs of the bit formation blocks are, therefore, functions formed by bits or waveforms in time domain.
    The 2430 frequency and time domain waveform provider can comprise the same number of bit-forming blocks as the frequency sub-bands in the frequency and time domain representation of the watermark data.
    According to another embodiment, the watermark design provider 2400 may comprise a weighted tuner.
    The weight tuner can also be called a psychoacoustic processing module.
    The weight tuner can be configured to tune the weight or a range of functions formed by bits corresponding to values of the 2410 frequency and time domain representation of the watermark data.
    A weight of a function formed by bits can be tuned in modification, as much energy as possible is attributed to a function formed by bits, however the watermark signal 2420 is still kept inaudible.
    The weight tuner can tune the weight in an iterative process for each function formed by bits corresponding to a value of the domain representation of. frequency and time 2410. Therefore, the weights of different functions formed by bits can vary. 2. METHOD FOR PROVIDING A WATERMARK SIGN
    Figure 25 shows a 2500 method of providing a watermark signal depending on a representation of the watermark data frequency and time domain.
    The method2500 comprises a first step 2510 of providing round forms of time domain for a plurality of frequency sub-bands, based on a representation of frequency domain and time of watermark data by mapping a determined value of the domain representation of frequency and time in a bit-forming function, in which a temporal extension of the bit-forming function is greater than the bit interval associated with the given value of the frequency domain representation and time, so that there is a temporal overlap between functions formed by bits provided for temporally subsequent values of representation of frequency domain and time of the same frequency range.
    A time domain waveform of a given frequency sub-band contains a plurality of bit-formed functions provided for time-subsequent values of the frequency domain and time representation of the same frequency sub-band. Method 2500 further comprises a step 2520 of combining waveforms provided for a plurality of frequencies to derive the watermark signal.
    The watermark signal can, for example, be a sum of the waveforms provided for the plurality of frequencies. Optionally, the 2500 method can comprise other steps corresponding to the characteristics of the equipment described above. 3. DESCRIPTION OF THE SYSTEM
    In the following, a system for watermark transmission will be described, comprising a watermark inserter and a watermark decoder.
    Naturally, the watermark insert and the watermark decoder can be used independently of each other. For the system description, a top-down approach is chosen here.
    First, it distinguishes between inter-encoder and decoder. Then, in sections 3.1 to 3.5, each processing block is described in detail.
    The basic structure of the system can be seen in Figures 1 and 2, which depict the encoder and dodecoder side, respectively. Figure 1 shows a schematic block diagram of a 100 watermark inserter.
    In the coding side, the watermark signal 101b is generated in the processing block 101 (also known as the watermark generator) of the binary data 101a and based on the information 104, 105 exchanged with the psychoacoustic processing module 102. Information Block 102 typically ensures that the watermark is inaudible.
    The watermark generated by the watermark generator 101 is then added to the audio signal 106. The watermark signal 107 can then be transmitted, stored or further processed.
    In the case of a multimedia file, for example, an audio and video file, an appropriate delay needs to be added for the video stream so as not to lose audio and video synchronization. In the case of a multi-channel audio signal, each channel is processed separately, as explained in this document.
    Processing blocks 101 (watermark generator) and 102 (psychoacoustic processing module) are explained in detail in Sections 3.1 and 3.2, respectively.
    The decoder side is depicted in Figure 2, which shows a schematic block diagram of a 200 watermark decoder. An audio signal with a watermark 200a, for example, recorded by a microphone, becomes available to the system200 .
    A first block 203, which is also designated as an analysis module, demodulates and transforms the data (for example, the audio signal with a watermark) into a time and frequency domain (thereby obtaining a frequency domain representation) time 204 of the audio signal with watermark 200a), passing it to the synchronization module 201, which analyzes the input signal 204 and performs a time synchronization, namely, determines the temporal alignment of the encoded data (for example, coded water related to the representation of frequency and time domain).
    This information (for example, the resulting synchronization information 205) is given to the watermark extractor 202, which decodes the data (and, consequently, provides the binary data 202a, which represents the data content of the d 'audio signal. water 200a). 3.1 THE WATERMARK GENERATOR 101
    The watermark generator 101 is depicted in details in Figure 3.
    The binary data (expressed as ± 1) to be hidden in the audio signal 106 is given to the watermark generator 101.
    Block 301 organizes data 101a in packets of length equal to Mp.
    Elevated bits are added (for example, affixed) for signaling purposes to each packet. Ms denotes your number.
    Its use will be explained in detail in Section 3.5. Note that, next, each packet of payload bits next to the high signaling bits is denoted as a message.
    Each message 301a, of length Nm = Ms + Mp, is delivered to processing block 302, the channel encoder, which is responsible for encoding the bits for protection against mistakes.
    A possible realization of this module consists of a convolutional encoder with an interleaver.
    The proportion of the convolutional encoder greatly influences the general degree of protection against errors in the watermark creation system.
    The interleaver, on the other hand, provides protection against meltdown explosions.
    The variation in interleaver operation can be limited to one message, but it could also be extended to more messages.
    Rc denotes the code ratio, for example, 1/4. The number of encoded debits for each message is Nra / Rc.
    The channel encoder provides, for example, an encoded binary message 302a. The next processing block, 303, performs a propagation in the frequency domain. In order to achieve sufficient signal-to-noise ratio, the information (for example, the information from the binary message 302a) is propagated and transmitted in carefully chosen sub-bands.
    Its exact frequency position is decided a priori and is known to both the encoder and the decoder. The details of choosing this important system parameter are given in Section 3.2.2. Frequency propagation is determined by the propagation sequence cf of size Nf X 1.
    The output 303a of block 303 consists of Nf bit streams, one for each subrange. The ith flow of bits is obtained by multiplying the input bit with the ith component of the propagation sequence cf.
    The simplest propagation consists of copying the bit stream to each output stream, namely, using a sequence of propagation of all. Block 304, which is also designated as a synchronization scheme inserter, adds a de-synchronization signal to the bit stream. .
    Strong synchronization is important, since the decoder does not know the time alignment of either the bits or the data structure, that is, when the message starts. The synchronization signal consists of N sequences of Nbit bits each.
    The strings are multiplied by element and periodically to the bit stream (or bit streams303a). For example, consider that a, b and c are the Ns = 3 synchronization sequences (also referred to as propagated synchronization sequences).
    Block 304 multiplies a to the first spread bit, b to the second spread bit and c to the third spread bit.
    For the following bits, the process is periodically repeated, namely, a to the fourth bit, b to the fifth bit and so on.
    Likewise, synchronization information and combined information 304a is obtained.
    Synchronization sequences (also referred to as propagated synchronization sequences) are carefully chosen to minimize the risk of false synchronization.
    More details are given in Section 3.4. Also, it should be noted that a sequence a, b, c, ... can be considered as a sequence of propagated synchronization sequences.
    Block 305 performs a propagation in the domain of. time.
    Each propagation bit in the input, namely a vector of length Nf, is repeated in the time domain Nt times. Similarly to the propagation in the frequency, we define a propagation sequence of size Nt X 1.
    The ith temporal repetition is multiplied with the ith component of ct. The operations of blocks 302 to 305 can be placed in mathematical terms, as follows.
    Consider m of size 1 X Nm = Rc a coded message, produced from 302. Asaida 303a (which can be considered as a representation of propagation information R) of blocks 303 is
    output 304a of block 304, which can be considered a representation of synchronization and combined information C, is
    where the product by Schur element denotes and
    Output 305a of 305 is
    Where
    denote the Kronecker product and transposition, respectively.
    Remember that the binary data is expressed as ± 1.The block 306 performs a different encoding of the bits.
    This step gives the system additional resistance against phase changes, due to mismatches of movement or location oscillators. More details on this will be given in Section 3.3. If b (i; j) is the bit for the nth frequency range and the j-th block of time at the input of block 306, the output bit
    It's
    At the beginning of the flow, that is, for j = 0, bdiff (i, j- 1) is set to 1.
    Block 307 performs real modulation, that is, generating the waveform of the watermark signal, depending on the binary information 360a given at its input. A more detailed scheme is given in Figure 4.
    In parallel inputs, 401 to 40N contain the bit streams for different sub-bands. Each bit of each sub-band flow is processed by a debit block (411 to 41Nf). The output of the bit-forming blocks are waveforms in the time domain. The waveform generated for the j-is-time block and the nth sub-band, denoted by you; j (t), based on the bdlff input nobit (i, j), is computed as follows
    where y (i; j) is a weighting factor provided by the psychoacoustic processing unit 102, Tb is the bit time interval, and gi (t) is the bit-forming function for the nth subrange.
    The bit-forming function is obtained from a baseband function
    frequency modulated with a cosine
    where fi is the central frequency of the nth sub-band and superscript T means the transmitter. The basic range functions can be different for each sub-range. If identical, a more efficient implementation in the decoder is possible.
    See Section 3.3 for more details. The bit formation for each bit is repeated in an iterative process controlled by the psicoacoustic processing module (102).
    Iterations are necessary to precisely adjust the y (i, j) weights, to assign as much energy as possible to the watermark, while keeping it inaudible. More details are given in Section 3.2.
    The complete waveform at the output of the 41i bit forming isofilter is
    The bit-forming baseband function
    it is not normally zero for a much longer time interval than TB, although the main energy is concentrated within the bit interval. An example can be seen in Figure 12a, where the same bit-forming baseband function is outlined by adjacent bits.
    In the figure, we have Tb = 40ms. The choice of Tbassim as the form of the function, affects the system considerably.
    In fact, longer symbols provide narrower frequency responses.
    This is particularly beneficial in reverberating environments.
    In fact, in these scenarios, the watermarked signal reaches the microphone through several propagation pathways, each characterized by a different propagation time.
    The resulting channel has strong frequency selectivity. Interpreted in the time domain, longer symbols are beneficial as echoes with a delay comparable to the constructive interference of bit interval production, which means that they increase the energy of the received signal.
    However, longer symbols also have some disadvantages; Larger overlaps could lead to inter-symbol interference (ISI) and are, of course, more • difficult to hide in the audio signal, so that the psychoacoustic processing module would allow less energy than for shorter symbols.
    The watermark signal is obtained by adding all the outputs of the bit formation filters
    3.2 THE PSYCHOACOUSTIC PROCESSING MODULE 102
    As shown in Figure 5, the psychoacoustic processing module 102 consists of 3 parts.
    The first step is an analysis module 501 that transforms the time audio signal into a time / frequency domain.
    This analysis module can perform parallel analyzes at different time / frequency resolutions.
    After the analysis module, the time / frequency data is transferred to the psychoacoustic model (PAM) 502, in which the masking limits for the watermark signal are calculated, according to the psychoacoustic considerations (see E. Zwicker H. Fasti , "Psychoacoustics Factsand models"). The masking limits indicate the amount of energy that can be hidden in the audio signal for each sub-band and time block.
    The last block in the psychoacoustic processing module 102 depicts the amplitude calculation module 503.
    This module determines the amplitude gains to be used for watermark signal nagering, so that the overcharging limits are satisfied, that is, the embedded energy is less than the energy defined by the masking limits. 3.2.1 TIME / FREQUENCY ANALYSIS 501
    Block 501 performs the time / frequency transformation of the audio signal by means of a rectified transformation.
    The best audio quality can be achieved when multiple time / frequency resolutions have been performed.
    An efficient realization of a rectified transformation is the short time Fourier transform (STFT), which is based on the fast Fourier transform (FFT) of the windowed time blocks.
    The length of the window determines the resolution of time / frequency, so that larger windows produce resolutions of shorter time and of higher frequency, and vice versa for shorter windows.
    The shape of the window, on the other hand, among other things, determines the loss of frequency.
    For the proposed system, we achieved an inaudible watermark by analyzing the data with the two different resolutions.
    A first filter bank is characterized by a hop size of Tb, that is, the length of the bit.
    The size is the time interval between two adjacent blocks of time.
    The length of the window is approximately Tb. Note that the shape of the window should not be the same as that used for bit formation and, in general, it should model the human auditory system.
    Several publications study this problem. The second filter bank applies a shorter window.
    The highest temporal resolution achieved is particularly important when embedding a watermark in speech, since its temporal structure is, in general, thinner than Tb. The sampling rate of the incoming audio signal is not important, as long as the enough to describe a watermark sign without aliasing.
    For example, if the broadest frequency component contained in the watermark signal is 6 kHz, then the sample rate of the time signals must be at least 12 kHz. 3.2.2 0 PSYCHOACOUSTIC MODEL 502
    The psychoacoustic model 502 has the task of determining the masking limits, that is, the amount of energy that can be hidden in the audio signal for each sub-band and time block, keeping the audio signal with watermark 10 indistinguishable from the original.
    The i th sub-band is defined between two limits, namely,
    and
    .
    The sub-bands are determined by defining Nf f (innx) r (nihi) center frequencies fi and leaving
    i for i = 2, 3, ..., Nf.
    An adequate choice for central frequencies is given by the 15 Bark scale proposed by Zwicker in 1961.
    Subbands become larger for higher center frequencies.
    A possible implementation of the system uses 9 sub-bands ranging from 1.5 to 6 kHz, properly arranged.
    The following processing steps are performed separately for each time / frequency resolution for each sub-band and each time block.
    Processing step 801 performs spectral smoothing.
    In fact, the tone elements as well as notches in the energy spectrum need to be smoothed out. This can be done in several ways.
    A shade measure can be computed and then used to trigger an adaptive smoothing filter.
    Alternatively, in a simpler implementation of this block, an average filter can be used. The median filter considers a vector of values and produces its median value.
    In a medium * type filter, the value corresponding to an amount other than 50% - can be chosen.
    The amplitude of the filter is defined in Hz and is applied as an average of non-linear movement that starts at the lower frequencies and ends at the highest possible frequency. The 801 operation is illustrated in Figure 7.
    The red curve is the result of smoothing. Once smoothing has been performed, the limits are computed by block 802, considering only the frequency masking.
    In this case, too, there are different possibilities. One way is to use the minimum for each sub-band to compute the Ei masking energy.
    That is, the equivalent energy of the signal that efficiently operates a masking. From this value, we can simply multiply a given scaling factor to obtain the masked energy Ji.
    These factors are different for the range and time / frequency resolution and are obtained through empirical psychoacoustic experiments. These steps are illustrated in Figure 8.
    In block 805, timeless masking is considered. In this case, different blocks of time for the same sub-range are analyzed.
    The masked energies Ji are modified according to an empirically derived post-masking profile. Let us consider two adjacent blocks of time, namely, k-1 and k.
    The corresponding masked energies are Ji (k-l) and Ji (k). The post-masking profile defines that, for example, the masking energy Ei can mask an energy Ji at moment k and α • Ji at moment k + 1.
    In this case, block 805 compares Ji (k) (the energy masked by the current time block) and »Ji (k + l) (the energy masked by the previous time block) and chooses the maximum. Post-masking profiles are available in the literature and were obtained through empirical psychoacoustic experiments.
    Note that for wide Tb, that is,> 20ms, post-masking is applied only to the time / frequency resolution with smaller time windows.
    In short, at the exit of block 805, we have the masking limits for each sub-range and time block obtained for two different time / frequency resolutions.
    Limits were obtained by considering both phenomena over frequency and time.
    In block 806, the limits for different time / frequency resolutions are mixed.
    For example, a possible implementation is that 806 considers all the limits corresponding to the time and frequency intervals, in which a bit is allocated, and chooses the minimum. 3.2.3 0 EXTENSION CALCULATION BLOCK 503
    See Figure 9. The entry of 503 are the limits505 of the psychoacoustic model 502, where all psychoacoustic motivated calculations are performed.
    In the amplitude calculator 503, additional computations are performed with the limits.
    First, a 901 amplitude mapping occurs. This block merely converts the masking limits (usually expressed as energies) into amplitudes that can be used to scale the bit formation function defined in Section 3.1. After that, amplitude adaptation block 902 is executed.
    This block adaptively adapts the amplitudes y (i, j) that are used to multiply the bit formation functions in the watermark generator 101, so that the masking limits are, in fact, met.
    In fact, as already discussed, the bit formation function normally extends over a period of time longer than TB.
    Therefore, multiplying the correct amplitude j) that meets the masking limit at point i, j does not necessarily meet the requirements at point i, j-1.
    This is particularly crucial in strong beginnings, as a pre-echo becomes audible.
    Another situation that needs to be avoided is an unreasonable overlap of the ends of different bits that could lead to an audible watermark.
    Therefore, block 902 analyzes the signal generated by the watermark generator, to verify that the limits are met.
    If not, the amplitudes y (i, j) are modified accordingly.
    This concludes the encoder side. The following sections deal with the processing steps performed on the receiver (also known as a watermark decoder). 3.3 0 ANALYSIS MODULE 203
    Analysis module 203 is the first step (or block) of the watermark extraction process.
    Its goal is to transform the audio signal with watermark 200a back into N bit streams
    (also designated with 204), one for registering spectral range i.
    They are further processed by the synchronization module 201 and the watermark extractor 202, as discussed in Sections 3.4 and 3.5, respectively.
    Notice that
    they are flexible bit streams, that is, they can, for example, have any real value and the rigid decision on the bit has not yet been made.
    This analysis module consists of three parts that are depicted in Figure 16: The analysis filter bank1600, the amplitude normalization block 1604 and the differential decoding 1608.f 3.3.1 ANALYSIS FILTER BANK 1600
    The watermarked audio signal is transformed into a time-frequency domain by the analysis filter bank1600, which is presented in detail in Figure 10a.
    The input to the filter bank is the audio signal with the receiving watermark (t).
    Its output is the complex coefficients
    for the iésimaramification or sub-band at the moment j.
    These values contain information about the amplitude and phase of the signal at the central frequency f £ and moment j-Tb.
    The filter bank 1600 consists of Nframifications, one for each spectral subrange i.
    Each branch is divided into an upper sub-branch for the phase component and a lower sub-branch for the sub-frame component of the sub-range i. Despite the modulation in the watermark generator and, thus, the audio signal with watermark are purely evaluated in real terms, the complex evaluated analysis of the signal in the receiver is necessary, because the rotations of the modulation constellation introduced by the channel and by the synchronization misalignments are not are known at the receiver.
    Next, we consider the iestimation of the filter bank. By combining the emphasis and quadrature sub-branching, we can define the complex baseline range signal
    like
    where * | indicates convolution and
    is the low-pass filter impulse response of the sub-range i receiver.
    Usually,
    (t) is equal to the base bit formation function of subband i in modulator 307, in order to meet the corresponding filter condition, but other impulse responses are also possible.
    In order to obtain the coefficients
    with the rate l = Tb, the output continues
    must be sampled. If the correct timing of the bits is known to the receiver, sampling with rate l = Tb would be sufficient.
    However, as bit synchronization is not yet known, sampling is carried out using the Nos / Tbc rate where Nos is the oversampling factor of the analysis filter bank. By choosing Nos sufficiently wide (for example, Nos = 4), we can guarantee that at least one sampling cycle is close enough to the optimal bit synchronization.
    The decision on the best oversampling layer is made during the synchronization process, thus, all oversampled data is maintained until that time. This process is described in Section 3.4. At the exit of the ith branch, we have the coefficients
    where j indicates the bit number or the moment and k indicates the oversampling position within that single bit, where k = 1; two ; . . . . , Nos.
    Figure 10b gives an exemplary overview of the location of the coefficients in the time and frequency plan.
    The oversampling factor is Nos = 2. The height and amplitude of the rectangles indicate respectively the range amplitude and the time interval of the part of the signal that is represented by the corresponding coefficient
    If the sub-range frequencies f ± are chosen as multiples of a given interval Δf, the analysis filter bank can be implemented efficiently using the Fast Fourier Transform (FFT). 3.3.2 AMPLITUDE STANDARDIZATION 1604
    Without losing generality and to simplify the description, we assume, next, that the bit synchronization is known and that Nos = 1. That is, we have complex coefficients at the input of the normalization block 1604.
    Since channel status information is not available at the receiver (ie, the propagation channel is unknown), an equal gain combination (EGC) scheme is used. Due to the time and frequency dispersive channel, the energy of the sent bit b ± (j) is found not only around the central frequency f ± and time moment j, but also in adjacent frequencies and moments.
    Therefore, for more precise weighting, additional coefficients are calculated at the frequencies
    and used to normalize the coefficient
    . If n = 1, we have, for example
    Normalization to n> 1 is a direct extension of the above formula.
    In the same way, we can also choose to normalize the flexible bits, when considering more than one moment. Normalization is performed for each subband i and each moment j.
    The real combination of EGC is done in the last stages of the extraction process. 3.3.3 DIFFERENTIAL DECODING 1608
    At the output of the differential decoding block 1608, we have the normalized complex amplitude coefficients
    which contain information about the phase of the signal components at the frequency ft and moment j.
    Since the bits are differentially encoded in the transmitter, the reverse operation must be performed here. The flexible bits
    are obtained, first, by calculating the difference in the phase of two consecutive coefficients and then obtaining the real part:
    This must be done separately for each sub-range, as the channel normally introduces different phase rotations in each sub-range. 3.4 THE SYNCHRONIZATION MODULE 201
    The task of the synchronization module is to find the time alignment of the watermark. The problem of synchronizing the decoder to the encoded data is twofold.
    In a first step, the analysis filter bank must be aligned to the encoded data, namely, the bit formation functions used in the synthesis in the modulator must be aligned to the filters used for analysis. This problem is illustrated in Figure 12a, where the analysis filters are identical to the synthetic ones.
    At the top, three bits are visible. For simplicity, waveforms for all three bits are not scaled.
    The time compensation between different bits is Tb. The bottom part illustrates the synchronization issue in the decoder: the filter can be applied at different times; however, only the position marked in red (curve 1299a) is correct and allows extracting the first bit with the best signal to SNR noise ratio and signal to SIR interference ratio.
    In fact, incorrect alignment would lead to a degradation of both SNR and SIR. We refer to that first alignment issue with bit synchronization ".
    Once bit synchronization has been achieved, the bits can be optimally extracted. * However, in order to correctly decode a message, it is * necessary to know at which bit a new message begins.
    This question is illustrated in Figure 12b and is referred to as message synchronization. In the stream of decoded bits, only the starting position marked in red (position 1299b) is correct and allows decoding the k-th message.
    First, we will only deal with message synchronization. The synchronization signature, as explained in Section 3.1, is made up of N strings in a predetermined order, which are embedded continuously and periodically in the watermark.
    The synchronization module is capable of recovering the time alignment of the synchronization sequences. Depending on the size Ns, we can distinguish between two modes of operation, which are depicted in Figure 12c and 12d, respectively.
    In the full message synchronization mode (Figure 12c), we have Ns = Nm / Rc. To simplify, in the figure, we assume Ns = Nm / Rc = 6 and without time propagation, that is, Nt = 1.
    The sync signature used for illustration purposes is shown below the messages. In reality, they are modulated depending on the encoded bits and frequency propagation sequences, as explained in Section 3.1. In this mode, the periodicity of the synchronization subscription is identical to that of messages.
    The synchronization module can, therefore, identify the beginning of each message, by finding the time alignment of the synchronization signature. We refer to the time positions in which a new synchronization subscription starts according to the synchronization beats.
    The synchronization strokes are then passed to the watermark extractor 202. * The second possible mode, the partial message synchronization mode * (Fig. 12d), is depicted in Figure 12d.
    In this case, we have Ns <Nm = Rc. In the figure, we consider Ns = 3, so that the three synchronization sequences are repeated twice for each message.
    Note that the periodicity of messages does not have to be a multiple of the periodicity of the synchronization subscription. In this operating mode, not all sync beats correspond to the beginning of a message.
    The synchronization module has no means of distinguishing between strikes and this task is given to the watermark extractor 202. The processing blocks of the synchronization module are depicted in Figures 11a and 11b.
    The synchronization module performs bit synchronization and message synchronization (complete or partial) at once, when analyzing the output of the synchronization signature correlator 1201. The data time / frequency domain 204 is provided by the analysis module .
    Since bit synchronization is not yet available, block 203 oversamples the data with the Nos factor, as described in Section 3.3.
    An illustration of the input data is given in Figure 12e. For this example, we consider Nos = 4, Nt - 2 and Ns = 3.
    In other words, the synchronization signature consists of 3 strings (denoted with a, b and c). The time propagation, in this case, with the propagation sequence ct = [1 1] T, simply repeats each bit twice in the time domain.
    The exact sync beats are denoted with arrows and correspond to the beginning of each sync subscription. The synchronization subscription period is Nt • Nos • Ns = Nsbl which is 2 • 4 • 3 = 24, for example.
    Due to the periodicity of the • synchronization signature, the synchronization signature correlator (1201) • arbitrarily divides the time axis into blocks, called search blocks, of size Nsbl, whose subscript represents the extension of the search block. Each search block must contain (typically contains) a synchronization beat, as shown in Figure 12f.
    Each of the Nsbl bits is a candidate synchronization beat.
    The task of block 1201 is to compute a probability measure for each candidate bit of each block. This information is then passed to block 1204 which computes the synchronization beats. 3.4.1 THE SYNCHRONIZATION SIGNATURE CORRELATOR
    For each of the Nsbl candidate synchronization positions, the synchronization signature correlator computes a probability measure, the latter is greater than more likely, that is, the time alignment (both bit and full or partial message synchronization) was found. The processing steps are shown in Figure 12g.
    In the same way, a sequence 1201a of the probability values, associated with different positional choices, can be obtained; Block 1301 performs the temporal propagation, that is, multiplies each Nt bits with the temporal propagation sequence ct and then adds them together. This is done for each of the Nf frequency sub-bands.
    Figure 13a presents an example. We consider the same parameters as those described in the previous section, namely, Nos = 4, Nt = 2 and Ns = 3.
    The candidate synchronization position is marked. From that bit, with 'Nos compensation, Nt • Ns are obtained by block 1301 and propagated in time with the sequence ct, so that Ns bits are left.
    In block 1302, the bits are multiplied by element with the Ns propagation sequences (see Figure 13b). In block 1303, the frequency propagation is performed, namely, each bit is multiplied with the propagation sequence cf and then added next to the frequency.
    At that point, if the synchronization positions are correct, we would have Ns bits decoded. Since the bits are not known to the receiver, block 1304 computes the probability measure when considering absolute values of Ns values and sums.
    The output of block 1304 is, in principle, a non-coherent correlator that aims at the synchronization subscription. In fact, when choosing a small Ns, namely, the partial message synchronization mode, it is possible to use synchronization sequences (for example, a, b, c) that are mutually orthogonal.
    In doing so, when the correlator is not correctly aligned with the signature, its output will be very small, ideally zero. When using the full message synchronization mode, it is advisable to use as many orthogonal synchronization sequences as possible and then create a signature by carefully choosing the order in which they are used.
    In this case, the same theory can be applied when looking for propagation sequences with good autocorrelation functions. When the correlator is only slightly misaligned, then the output of the correlator will not be zero, even in the ideal case, • but, in any case, it will be less compared to the perfect alignment, since the analysis filters cannot capture the signal energy of ideal way. 3.4.2 SYNCHRONIZING BEATS COMPUTER 1204
    This block analyzes the output of the synchronization signature correlator to decide where the synchronization positions are.
    Since the system is quite resistant to misalignments of up to Tb / 4 and Tb is normally considered to be about 40 ms, it is possible to integrate the 1201 output over time to achieve a more stable synchronization. A possible implementation of this is given by an IIR filter applied over time with an exponentially decay impulse response.
    Alternatively, a traditional FIR traditional movement filter can be applied. Once the measurement has been performed, a second correlation across different Nt'Ns is performed ("different positioning choice").
    In fact, we want to explore the information that the autocorrelation function of the synchronization function is known for. This corresponds to a Maximum Probability estimator.
    The idea is presented in Figure 13c. The curve shows the exit of block 1201 after the temporal integration.
    One possibility of determining the timing beat is simply to find the maximum of this function. In Figure 13d, we see the same function (in black) filtered with the autocorrelation function of the synchronization signature.
    The resulting function is plotted in red. In this case, the maximum is more pronounced and gives us apposition of the synchronization beat.
    The two methods are quite similar for high SNR, but the second method performs much better in lower SNR regimes. Once the sync beats have been found, they are passed to the watermark extractor 202 that decodes the data.
    In some embodiments, in order to obtain a strong synchronization signal, synchronization is performed in the partial message synchronization mode with short unsynchronization signatures. For this reason, many decryptions have to be made, increasing the risk of false positive message detections.
    To avoid this, in some realizations, signal sequences with a lower bit rate may be inserted as a consequence. This approach is a solution to the problem that arises from a synchronization signature that is shorter than the message, which is already dealt with in the discussion above of the enhanced synchronization.
    In this case, the decoder does not know where a new message starts and tries to decode at different points of synchronization. To distinguish between legitimate and false positive messages, in some embodiments, a signal word is used (that is, the payload is sacrificed to embed a known control sequence).
    In some embodiments, a plausibility check is used (alternatively or additionally) to distinguish between legitimate and false positive messages. 3.5 THE WATERFALL EXTRACTOR 2 02
    The parts that make up the watermark extractor 202 are depicted in Figure 14. It has two entrances, known, 204 and 205 of blocks 203 and 201, respectively.
    The synchronization module 201 (see Section 3.4) provides synchronization time stamps, that is, the positions in the time domain at which a candidate message begins.
    More details on this subject are given in Section 3.4.
    The analysis filter bank block203, on the other hand, provides the data in the time / frequency domain ready to be decoded. The first processing step, the data selection block 1501, selects from the entry 204 the part identified as a message candidate be decoded.
    Figure 15b presents this procedure graphically. The 204 entry consists of Nf real value flows.
    Since the time alignment is not known to the decoder a priori, the analysis block 203 performs a frequency analysis with a rate greater than 1 / Tb Hz (oversampling). In Figure 15b, we use an oversampling factor of 4, namely, 4 vectors of size Nf X 1 are produced every Tb seconds.
    When the synchronization block 201 identifies a candidate message, it releases a timestamp 205 that indicates the start point of a candidate message. Selection block 1501 selects the information needed for decoding, namely a matrix of size Nf X Nm / Rc.
    This matrix 1501a is given to block 1502 for further processing. Blocks 1502, 1503 and 1504 perform the same operations as blocks 1301, 1302 and 1303 explained in Section 3.4. An alternative embodiment of the invention consists of avoiding the computations made in 1502-1504, by leaving that the synchronization module also releases the data to be decoded.
    Conceptually, this is a detail. From the point of view of implementation, it is only a matter of how the warehouses are carried out.
    In general, redoing the computations allows for smaller storages. The 1505 channel decoder performs the reverse operation of block 302.
    If the channel encoder, in a possible realization of this module, consisted of a convolutional encoder together with an interleaver, then the channel decoder would perform deinterlacing and convolutional decoding, for example, with the well-known Viterbi algorithm.
    In said block, we have Nmbits, that is, a candidate message. Block 1506, the signaling block and openness, decides whether the incoming candidate message is indeed a message or not.
    To do this, different strategies are possible. The basic idea is to use a word deinalization (like a CRC string) to distinguish between real and false inter-messages.
    However, this reduces the number of discussions available as a payload. Alternatively, we can use plausibility checks.
    If messages, for example, contain a time stamp, consecutive messages must have consecutive time stamps. If a decoded message has a time stamp that is not the correct order, we can discard it.
    When a message is correctly detected, the system can choose to apply the anticipation and / or agreement mechanisms. We assume that debit and message synchronizations have been achieved.
    Assuming that the user is not zapping, the system "anticipates" time and tries to decode previous messages (if not already decoded) using the same synchronization point (recall approach).
    This is particularly useful when the system starts. In addition, in poor conditions, two messages could be considered to achieve synchronization.
    In that case, the first message has no chance. With the option of 5 advance, we can save "good" messages that were not received just due to bad synchronization.
    The anticipation is the same, but it works in the future. If we have a message, we now know where the next message should be and we can try to decode it anyway. 3.6. SYNCHRONIZATION DETAILS
    For coding a payload, for example, a Viterbi algorithm can be used.
    Figure 18a is a graphical representation of an 1810 payload, a Viterbi 1820 closure sequence, a Viterbi 1830 encoded payload 15 and an 1840 repeat encoded version of the Viterbi encoded payload.
    For example, the payload extension can be 34 bits and the Viterbi termination sequence can comprise 6 bits.
    If, for example, a Viterbi code rate of 1/7 can be used, the payload encoded by Viterbi can comprise (34 + 6) * 7 = 280 bits.
    In addition, when using a 1/2 repetition encoding, the 1840 repetition encoded version of the Viterbi 1830 encoded payload can comprise 280 * 2 = 560 bits.
    In this example, considering a bit time interval of 42.66 ms, the message length would be 23.9 s.
    The signal can be embedded with, for example, 9 subchargers (for example, placed according to the important bands) from 1.5 to 6 kHz, as indicated by the frequency spectrum shown in Figure 18b.
    Alternatively, another number of subchargers (eg 4, 6, 12, 15 or a number between 2 and '20) within a frequency range between 0 and 20 kHz can be used *.
    Figure 19 presents a schematic illustration of the basic concept 1900 for synchronization, also called ABC synchronization.
    A schematic illustration of unencoded messages 1910, an encoded message 1920 and a synchronization sequence (sync sequence) 1930 is shown, as well as the application of synchronization to several 1920 messages, one after the other. The sync sequence or sync sequence. mentioned in connection with the explanation of this synchronization concept (shown in Figures 19 to 23) can be the same as the synchronization subscription mentioned above.
    In addition, Figure 20 presents a schematic illustration of the synchronization found when correlating with the sync sequence.
    If the 1930 sync sequence is less than the message, more than one 1940 sync point (or alignment time block) can be found within a single message.
    In the example shown in Figure 20, four synchronization points are found within each message.
    Therefore, for each synchronization found, a Viterbi decoder (a Viterbi decoding sequence) can be started.
    In this way, for each 1940 synchronization point, a 2110 message can be obtained, as shown in Figure 21.
    Based on these messages, the actual 2210 messages can be identified through a CRC sequence (cyclical redundancy check sequence) and / or a plausibility check, as shown in Figure 22.
    CRC detection (cyclic redundancy check detection) can use a known sequence to identify the actual false positive messages.
    Figure 23 5 shows an example for a CRC sequence added to the end of a payload.
    The probability of a false positive (a message generated based on a wrong sync point) may depend on the length of the CRC sequence and the number of 10 Viterbi decoders (number of sync points within a single message) initiated.
    To increase the length of the payload without increasing the probability of a false positive, a plausibility can be explored (plausibility test) or the length of the synchronization sequence (synchronization signature) can be increased. 4. CONCEPTS AND ADVANTAGES
    Below, some aspects of the system discussed above will be described, which are considered innovative. Also, the relationship of these aspects to those of prior art technologies 20 will be discussed. 4.1 CONTINUOUS SYNCHRONIZATION
    Some achievements allow for continuous synchronization. The synchronization signal, which we denote as a synchronization signature, is embedded continuously and in parallel to the data 25 by means of multiplication with sequences (also known as propagated synchronization sequences) known both at the transmission and at the receiving side.
    Some conventional systems use special symbols (different from those used for data), while * some embodiments, according to the invention, do not use these * special symbols. Other classic methods consist of embedding a known bit sequence (preamble) multiplexed in time 5 with the data or embedding a signal multiplexed in frequency with the data.
    However, it was found that the use of dedicated sub-bands for synchronization is not desired, since the channel could have notches in these frequencies, making synchronization 10 unreliable. Compared to other methods, in which a preamble or a special symbol is multiplexed in time with the data, the method described here is more advantageous, since the method described here allows to track changes in synchronization (due, for example, to movement) continuously.
    In addition, the energy of the watermark signal is unchanged (for example, by the multiplicative introduction of the watermark in the representation of propagation information) and the synchronization can be designated independent of the psychoacoustic model and data rate.
    The time extension of the synchronization subscription 20, which determines the resistance of the synchronization, can be designated, at will, completely independent of the data rate.
    Another classic method consists of embedding a multiplexed synchronization sequence in code with the data.25 When compared to this classic method, the advantage of the method described here is that the energy of the data does not represent an interference factor in the computation of the correlation, bringing more resistance.
    In addition, when using code multiplexing, the number of orthogonal sequences available for synchronization is reduced as much as necessary for the data.
    To summarize, the synchronization approach described here brings with it a wide number of advantages over conventional concepts. However, in some embodiments, according to the invention, a different synchronization concept can be applied. 4.2. 2D PROPAGATION
    Some realizations of the proposed system carry out propagation in both the time and frequency domains, that is, a two-dimensional propagation (abbreviated as 2D propagation).
    This has been found to be advantageous in relation to 1D systems, since the bit error can be further reduced by adding redundancy, for example, in the time domain. However, in some embodiments, according to the invention, a different propagation concept can be applied. 4.3. DIFFERENTIAL ENCODING AND DIFFERENTIAL DECODING
    In some embodiments, according to the invention, greater resistance against mismatch of movement and frequency of local oscillators (when compared to conventional systems) is produced by differential modulation.
    It was found that, in fact, the Doppler effect (movement) and the frequency non-correspondence lead to a rotation of the BPSK constellation (in other words, a rotation in the complex plane of the bits).
    In some embodiments, the damaging effects of disarming the BPSK constellation (or any other properly modulated constellation) are avoided when using differential encoding or differential decoding.
    However, in some embodiments, according to the invention, a different coding concept or decoding concept may be applied.
    Also, in some cases, differential encoding can be omitted. 4.4. BIT FORMATION
    In some embodiments, according to the invention, the formation of bits brings a significant improvement in the performance of the system, since the reliability of the detection can be increased using a filter adapted to the formation of bits.
    According to some achievements, the use of bit formation in relation to watermark creation brings with it improved reliability of the watermark creation process.15 It was found that particularly good results can be obtained if the function of bit formation is greater than the bit range.
    However, in some embodiments, according to the invention, a different bit-forming concept can be applied. Also, in some cases, bit formation can be omitted. 4.5. INTERACTIVE SYNTHESIS BETWEEN THE PSYCHOACOUSTIC MODEL (PAM) AND FILTER BANK (FB)
    In some embodiments, the psychoacoustic model interacts with the modulator to precisely adjust the amplitudes 25 that multiply the bits.
    However, in some other realizations, this interaction can be omitted. 4.6. ASPECTS OF ANTICIPATION AND RECALL
    In some embodiments, so-called "Anticipation" and "Recall" approaches are applied.
    In the following, these concepts will be briefly summarized. When a message is correctly decoded, it is assumed that synchronization has been achieved.
    Assuming that the user is not zapping, in some achievements, an anticipation of time is performed and an attempt is made to decode the previous messages (if not already decoded), using the same point of synchronization (recall approach).
    This is particularly useful when the system starts up. In poor condition, two messages could be considered to achieve synchronization.
    In this case, the first message has no chance in conventional systems.
    With the recall option, which is used in some embodiments of the invention, it is possible to save (or decode) "good" messages that were not received only due to back synchronization.
    The anticipation is the same, but it works in the future. If I have a message, now I know where my next message should be and I can try to decode it anyway.
    In the same way, the overlay messages can be decoded. However, in some embodiments, according to the invention, the anticipation aspect and / or the recall aspect can be omitted. 4.7. INCREASED SYNCHRONIZATION RESISTANCE
    In some embodiments, in order to obtain a strong synchronization signal, synchronization is carried out in partial message synchronization mode with short synchronization signatures.
    For this reason, many decryptions have to be made, increasing the risk of detecting false, positive messages.
    To avoid this, in some embodiments, signaling sequences can be inserted in messages with a lower bit rate, as a consequence. However, in some embodiments, according to the invention, a different concept to improve synchronization resistance can be applied.
    Also, in some cases, the use of any concepts to increase synchronization resistance can be omitted. 4.8. OTHER IMPROVEMENTS
    Below, some other general improvements of the system described above in relation to the prior art will be presented and discussed: 1. minor computational complexity2. better audio quality due to the best psychoacoustic model3. more resistance in reverberant environments due to narrow band multi-charger signals4. an SNR estimate is avoided in some realizations.
    This allows better resistance, especially in low SNR regimes. Some embodiments, according to the invention, are better than conventional systems, which use very narrow range amplitudes, for example, 8 Hz due to the following reasons: 1 . 8 Hz bandwidths (or a similar very narrow bandwidth) require very long time symbols because the psychoacoustic model allows very little energy to make it inaudible; 2. 8 Hz (or a very narrow bandwidth similar) becomes sensitive in relation to Doppler spectra that vary over time.
    Likewise, this narrowband system is typically not good enough if implemented, for example, in a watch.
    Some realizations, according to the invention, are better than other technologies due to the following reasons: 1. Techniques that insert an echo completely fail in reverberant environments. On the contrary, in some embodiments of the invention, the introduction of an echo is avoided.2. Techniques that use only time propagation have a longer message duration compared to the realizations of the system described above, in which a two-dimensional propagation, for example, both in time and in frequency, is used.
    Some achievements, according to the invention, are better than the system described in document DE 196 40 814, since one or more of the following disadvantages of the system according to said document are overcome: • the complexity in the decoder, according to the document DE 196 40 814 is very high, a 2N length filter with N = 128 is used • the system, according to DE 19640 814, comprises a long message duration • in the system, according to DE 19640 814, a propagation only in the time domain with relatively high prepayment gain (eg 128) • in the system, according to document DE 196 40 814, the signal is generated in the time domain, transformed to the spectral domain, weighted, transformed back to time domain and superimposed on the audio, which makes the system very complex. 5. APPLICATIONS
    The invention comprises a method for modifying an audio signal in order to hide digital data and a corresponding decoder capable of recovering that information, while the perceived quality of the modified audio signal remains indistinguishable from that of the original.
    Examples of possible applications of the invention are given below: 1. Transmission monitoring: information containing a watermark, for example, about the station and time is hidden in the audio signal of radio or television programs. The decoders, incorporated in small devices used by test subjects, are able to recover the watermark and, thus, collect valuable information for advertising agencies, as well as, who watched the program and when. Audit: a watermark can be hidden, for example, in advertisements. By automatically monitoring astransmissions from a given station, then, it is possible to know when exactly the advertisement was broadcast.
    Similarly, it is possible to retrieve statistical information about the programming schedules of different radios, for example, how often a particular musical track is played, etc.3. Embedding of metadata: the proposed method can be used to hide digital information about the track of the song or program, for example, the name and author of the track or the duration of the program etc. „ 6. IMPLEMENTATION ALTERNATIVES
    Although some aspects have been described in the context of an equipment, it is clear that these aspects 5 also represent a description of the corresponding method, where a block or device corresponds to a method step or an aspect of a method step.
    Similarly, the aspects described in the context of a method step also represent a description of a corresponding block or item or aspect of a corresponding equipment 10.
    Some or all steps of the method can be performed by (or using) hardware equipment, such as a microprocessor, a programmable computer or an electronic circuit.
    In some embodiments, some or more of the most important method steps can be performed by this equipment. The coded watermark signal of the invention or an audio signal in which the watermark signal is embedded can be stored on a digital storage medium or it can be transmitted on a transmission medium, such as a wireless transmission medium or a wired transmission medium, such as the Internet.
    Depending on certain implementation needs, the realizations of the invention can be implemented in hardware or in software.
    The implementation can be carried out using a digital storage medium, for example, a floppy disk, a DVD, a Blu-Ray, a CD, a ROM, a PROM, an EPROM, an EEPROM, or a FLASH memory, which has signals legible control devices electronically stored in it, which cooperate (or are capable of cooperating) with a programmable computer system, whenever the respective method is carried out.
    Therefore, the digital storage medium can be computer readable. Some embodiments, according to the invention, comprise a data loader that has electronically controllable signals, which are able to cooperate with a programmable computer system, so that one of the methods aquidescritos is carried out.
    In general, the achievements of the present invention can be implemented as a computer program product with a program code, the program code being operated to perform one of the methods when the computer program product runs on a computer.
    The program code can, for example, be stored in a machine-readable charger.
    Other achievements include the computer program to perform one of the methods described here, stored in a machine-readable charger.
    In other words, an embodiment of the invention method is, therefore, a computer program having a program code to perform one of the methods described here when the computer program runs on a computer.
    A further embodiment of the methods of the invention is, therefore, a data loader (or a digital storage medium or a computer-readable medium) comprising, written on it, the computer program for carrying out one of the methods described above. An additional embodiment of the method of the invention is therefore, a data stream or a sequence of signals representing the computer program to perform one of the methods described here.
    The data stream or signal sequence can, for example, be configured to be transferred via a data communication connection, for example, via the Internet. An additional realization comprises processing means, for example, a computer or a device programmable logic, configured or adapted to perform one of the methods described here.
    An additional realization comprises a computer with the computer program installed to perform one of the methods described here. In some embodiments, a programmable logic device (for example, a programmable field gate matrix) can be used to perform some or all of the method's functionality. described here.
    In some embodiments, a programmable field gate array can operate with a microprocessor in order to perform one of the methods described here. In general, the methods are preferably performed by any hardware equipment.
    The embodiments described above are merely illustrative for the principles of the present invention. It is understood that modifications and variations of the provisions and details described here will be apparent to those skilled in the art.
    It is intended, therefore, to be limited only by the scope of the impending patent claims and not by the specific details presented in the description and explanation of the achievements here.
    权利要求:
    Claims (10)
    [0001]
    1. WATERPROOF SIGNAL PROVIDER (2400; 307) TO PROVIDE A WATERPROOF SIGN (2420, wms (t); 307a; 101b) DEPENDING ON A FREQUENCY AND TIME DOMAIN REPRESENTATION (2410; bdiff (i, j); 401-40NJ BRAND DATA, where the frequency and time domain representation (2410; bdiff (z, j); 401- 40Nf) is characterized by comprises values associated with sub-bands of frequency (i) and bit intervals (j), the watermark signal provider (2400; 307) comprising: a frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf ) configured to provide time domain waveforms (2440; si (t)) for a plurality of frequency sub-bands (i), based on the representation of frequency and time domain (2410; bdiff (i, j); 401 - 40Nf) of the watermark data, in which the waveform provider of frequency and time domain (2430; 411-41Nf, 421-42Nf) is configured to map a certain value (bdiff (i, j) ) of the representation of the frequency and time domain (2410; bdiff (i, j); 401- 40Nf) in a bit-forming function (gi (t)), in which a time extension of the bit-forming function (gi (t)) is greater than the bit interval (j) associated with a given value ( bdiff (i, j)) of the frequency and time domain representation (2410; bdiff (i, j); 401- 40Nf), so that there is a time overlap between functions formed by bits (gi (t)) provided for temporally subsequent values of the frequency and time domain representation (2410; bdiff (i, j); 401-40Nf) of the same frequency sub-range (i); and that the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) is further configured so that a time domain waveform (2440, sit)) of a given subband frequency (i) contains a plurality of functions formed by bits (sij (t)) provided for temporally subsequent values of the frequency and time domain representation (2410; bdiff (i, j); 401-40Nf) of the same frequency range (i); and a time domain waveform combiner (2460) for combining the time domain waveforms (2440, sij (t)) provided for a plurality of frequencies (i) from the frequency and time domain provider (2430; 411-41Nf, 421- 42Nf) to derive the watermark signal (2420, wms (t); 307a; 101b); where the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) is configured so that a function formed by bits (si, j (t)) provided for a given value bdiff (i, j) of the frequency and time domain representation (2410; bdiff (i, j) , 401-40Nf) is superimposed by a function formed by bits of a temporally previous value (bdiff (i, j-1)) of the same frequency sub-band (i) as the given value (bdiff (i, j)) of the representation of frequency and time domain (2410; bdiff (i, j); 401-40Nf) and a function formed by bits (si, j + 1 (t)) of a temporally following value (bt J + 1 (t)) of the same frequency sub-range (i) as the given v alue (Zz; j (í)) of the frequency and time domain representation (2410; bdiB (i, j); 401-40Nf), so that a time domain waveform (2440, 5; (í)) provided by the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) contains a overlap between at least three functions formed by bits (si j (t)) temporally subsequent of the same frequency sub-range (i).
    [0002]
    2. WATERPROOF SIGNAL PROVIDER (2400; 307), according to claim 1, characterized in that the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) is configured so that a function formed by bits () provided for a given value báiB (i, j) of the representation of the domain of frequency and time (2410; báiB (i, j), 401-40Nf) is superimposed by a function formed by bits (st) of a previously timed value (biiB (i, j -1)) of the same frequency sub-range (i) as the given value (báiB (z, j)) of the frequency and time domain representation (2410; bdiB ( i, j); 401-40Nf) and a function formed by bits (sij + 1 (t)) of a temporally following value (bij + 1 (t)) of the same frequency sub-range (i) as the given value ( Zzjj (í)) of the frequency and time domain representation (2410; & diff (z, j); 401-40Nf), so that a time domain waveform (2440, sz (í)) provided by the provider domain waveform frequency and time (2430; 411-41Nf, 421-42Nf) contains an overlap between at least three functions formed by bits () temporally subsequent of the same frequency sub-band (i) •
    [0003]
    3. WATERMARK SIGNAL PROVIDER (2400; 307), according to claim 1, characterized in that the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) is configured so that a time extension of a bit-forming function (2450, g; (z)) is a time band, in which the bit-forming function (2450, gt (t)) comprises non-zero values, and in that the time band is at least three bit intervals (j) in length.
    [0004]
    4. WATERMARK SIGNAL PROVIDER (2400; 307), according to claim 1, characterized in that the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) is configured so that a bit-forming function (2450, g; (í)) is based on a amplitude-modulated periodic signal, where an amplitude modulation of the amplitude-modulated periodic signal is based on a base band function (gf (t)), in which the time extension of the bit forming function (2450, g, (í)) is based on the base band function (gf (/)); eem that i designates an index for a frequency sub-band, T designates transmitter, and t designates a time variable.
    [0005]
    5. WATERMARK SIGNAL PROVIDER (2400; 307), according to claim 4, characterized in that the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) is configured so that the base band function (gT (t)) is identical for a plurality of frequency sub bands (i) of the frequency and time domain representation (2 410; & diff (ú. /); 401-40NJ.
    [0006]
    6. WATERMARK SIGNAL PROVIDER (2400; 307), according to claim 4, characterized by a periodic part of the bit-forming function (2450, g (í)) is based on a cosine function, of so that gt (0 = gf (0 • cos (2π //), where cos is a cosine function and fj is a central frequency of a corresponding frequency sub-band (i) of the bit-forming function (2450, g, ( O ) •
    [0007]
    7. WATERPROOF SIGNAL PROVIDER (2400; 307), according to claim 1, characterized in that it also comprises a weight tuner (102) to tune a weight (105, / (*, /)) of a function formed by bits (^ (í)) provided for a given value (bdiff (i, j)) of the frequency and time domain representation (2410; bdiff (i, j); 401-40NJ, so that = where the weight tuner (102) is configured to tune the weight (105,) so that a power of the function formed by bits () is maximized in relation to inaudibility.
    [0008]
    8. WATERPROOF SIGNAL PROVIDER (2400; 307), according to claim 1, characterized in that the frequency and time domain waveform provider (2430; 411-41Nf, 421-42Nf) is configured so that a time domain waveform (2440, 5; (í)) of a given frequency sub-band (i) is a sum of all bit-formed functions ( j (0) of the given frequency sub-band (i), so that
    [0009]
    9. WATERPROOF SIGNAL PROVIDER (2400; 307), according to claim 1, characterized in that the time domain waveform combiner (2460) is configured so that the watermark signal (2420, wms (t); 307a; 101b) is a sum of the waveforms provided (2440, 5; (z)) for the plurality of frequency sub-bands (i), so that wms (t) =, y . (z) .i
    [0010]
    10. METHOD (2500) TO PROVIDE A MARCAD'AGUA SIGN (2420, wms (t); 307a; 101b) IN DEPENDENCE OF A FREQUENCY AND TIME DOMAIN PRESENTATION (2410; bdiB (z, j); 401- 40Nf) DATA BRAND DATA, in which the frequency and time domain representation (2410; & diff 401-40Nf) is characterized by comprising values associated with frequency sub-bands (i) and bit intervals (j), the method ( 2500) comprising: provision (2510) of time domain waveforms (2440, s ^ t)) for a plurality of frequency sub-bands (i), based on the representation of frequency and time domain (2410; bdiB ( i, j); 401-40Nf) of the watermark data, by mapping a certain value (bdiB (ij9) of the frequency time domain representation (2410; bdiB (i, j); 401-40Nf) in a bit-forming function (2450, g; (t)), where a time extension of the bit-forming function (2450, g; (í)) is greater than the bit interval (j) associated with a given value ( èdiff (z, j)) from repr presence of frequency and time domain (2410; bdiB (i, j); 401-40Nf), so that there is a temporal overlap between functions formed by bits () provided for temporally subsequent values of the representation of frequency domain and time (2410; báiB (i, j); 401- 40Nf) of the same frequency sub-range (i), and so that a time domain waveform (2440, 5, (0) of a certain frequency range (i) contains a plurality of functions formed by bits (5i7 (í)) provided for time values subsequent representations of the frequency and time domain representation (2410; bdiB (i, j); 401-40Nf) of the same frequency range (i); and combination (2520) of the time domain waveforms (2440, 5, ( í)> provided for the plurality of frequencies to derive the watermark signal (2420, wms (t); 307a; 101b); where a function formed by bits (5; j (í)) provided for a given bdiB (i, j) value of the frequency and time domain representation (2410; bdiB (i, j), 401-40Nf) is superimposed by a function formed by bits (s {) of a val or temporally anterior (bdiB (z, j -1)) of the same frequency sub-range (i) as the given value (èdiff (z, j)) representation of frequency and time domain (2410; bdiB (i, j); 401-40NJ and by a function formed by bits (sij + 1 (t)) of a temporally following value (bij + 1 (t)) of the same frequency sub-range (i) as the given value (& ij (0) of the representation frequency and time domain (2410; bdiB (i, j); 401-40Nf), so that the time domain (2440, 5; (z)) waveform provided contains an overlap between at least three functions formed by bits () temporally subsequent of the same frequency sub-range (i).
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    同族专利:
    公开号 | 公开日
    EP2539891A1|2013-01-02|
    RU2012140871A|2014-04-10|
    KR101401174B1|2014-05-29|
    CN102859585A|2013-01-02|
    CA2790973A1|2011-09-01|
    EP2539891B8|2014-04-23|
    EP2539891B1|2014-01-15|
    CA2790973C|2016-05-31|
    PL2539891T3|2014-07-31|
    SG183486A1|2012-10-30|
    CN102859585B|2014-07-02|
    US9214159B2|2015-12-15|
    KR20120128149A|2012-11-26|
    JP5426781B2|2014-02-26|
    WO2011104283A1|2011-09-01|
    HK1180445A1|2013-10-18|
    AU2011219796B2|2014-08-21|
    MX2012009788A|2012-11-23|
    ZA201206357B|2013-05-29|
    US20130261778A1|2013-10-03|
    BR112012021533A2|2017-07-04|
    MY161513A|2017-04-28|
    ES2452920T3|2014-04-03|
    JP2013520696A|2013-06-06|
    EP2362382A1|2011-08-31|
    AU2011219796A1|2012-09-27|
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    法律状态:
    2019-01-08| B06F| Objections, documents and/or translations needed after an examination request according [chapter 6.6 patent gazette]|
    2019-09-10| B06U| Preliminary requirement: requests with searches performed by other patent offices: procedure suspended [chapter 6.21 patent gazette]|
    2020-04-14| B09A| Decision: intention to grant [chapter 9.1 patent gazette]|
    2020-11-10| B16A| Patent or certificate of addition of invention granted [chapter 16.1 patent gazette]|Free format text: PRAZO DE VALIDADE: 20 (VINTE) ANOS CONTADOS A PARTIR DE 23/02/2011, OBSERVADAS AS CONDICOES LEGAIS. |
    优先权:
    申请号 | 申请日 | 专利标题
    EP10154948A|EP2362382A1|2010-02-26|2010-02-26|Watermark signal provider and method for providing a watermark signal|
    EP10154948.3|2010-02-26|
    PCT/EP2011/052694|WO2011104283A1|2010-02-26|2011-02-23|Watermark signal provider and method for providing a watermark signal|
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